60 Ampere BLDC. PCB design and routing

[DrGeoff]

I've seen the use of solid copper rods soldered onto the pads in some SMPS designs that required higher current rails.
They looked like 2mm diameter copper rods and were soldered directly onto the copper traces, the entire length of the trace (which was straight).

[satyamfifa]

Hey Thanks for your reply,

I according to my Gate-Driver datasheet calculation I should be able to drive a MOSFET with Qg = 333nC with IVCP of 15mA at 45kHz PWM switching frequency, does higher switching frequency offer any significant advantage?

The whole point of using DRV8323 is to able to do Field Oriented Control with less hassle, that is to make a board which is compatible to TI-InstaSPIN, unfortunately I have not seen a single example of a TI driver supporting instaspin with parallel MOSFETs, and best example I was was a 50 A E-bike with TI-InstaSPIN again with no parallel MOSFETs, currently I don't see reason why it may not be doable as long as stray inductance is kept under a certain limit. 

[rx8pilot]

Are you familiar with switching losses in power converters?

It is a big a thick topic - but for what you are doing, important.

[satyamfifa]

Nope I don't, although at this moment I wish I did, I thought switching losses low rise and fall time can be compensated with lower PWM frequency

[rx8pilot]

Switching losses happen during the transition from off-on and on-off. When the MOSFET is off - it dissipates no power. When it is on it generally dissipates very little power. During the time it takes to get from one state to the other is very lossy and a lot of power is dissipated - essentially this is when the MOSFET is behaving like a resistor.

The PWM frequency determines how many of these switching cycles you go through per unit of time - you simply multiply the losses from each switch cycle by the number of cycles to get the total loss. Therefore, lower frequency PWM will lower losses at the expense of needing larger inductors and capacitors. Once you slow the PWM down to the audible range - you will hear it squealing or whining and the inductors and capacitors will be really big.

In very general terms - the goal is to charge and discharge the gate as fast as possible. The rise time of the gate is determined by the gate charge the current available to fill it up. If your design is very low inductance - (meaning a really good PCB layout) - you can charge the gate very fast and minimize the resistive switching losses. If you have a considerable amount of parasitic inductance in the devices and PCB you will start to see overshoots and ringing that can cause other problems. It is common to put a resistor on the gate trace to have control over the charge time to compensate for the PCB.

I tend to design converters with the highest switching speed I can while hitting the other design targets - efficiency, size, etc... For high current, parallel FETs or interleaved converters deliver high-efficiency and high current while still being small.

[satyamfifa]

https://www.eevblog.com/forum/projects/lost-trying-to-calculate-switching-time-of-power-mosfet/

[satyamfifa]

Hey All,

Thanks for the replies, I think I have finally made up my mind about what I am going to do:

1) Use DirectFET MOSFET's to keep the stray inductance as low as possible as the main issue with no parallel MOSFET is that, and try not destroy MOSFETs by putting them through avalanche.

2) Go go for low Rds(on) MOSFET instead of fast switcihng MOSFET (which takes me right back to IRF7749/AUIRF7749), as swtiching characteristics are unpredicatble anyways, use a PWM frequecny of about 45kHz to minizime the switching losses and keep the MOSFET in conducation region.

3) High switching speed isn't desirable anyways again to avoid high avalanche peaks as suggested my Tim in his previous posts, therefor 1A source of DRV8323 should be enough.

4) Place large capacitors as close to MOSFETs as possible for the low PWM frequency. Again something I saw in Direct Drive ESC example.

5) Design PCB with shortest trace length, provide ample amount of area for heat dissipation.

6) Try to cover as much area of PCB with heatsink as possible, again as suggested by Tim.

7) I am gonna use double sided track both top and bottom (90 micrometer each) to dissipate heat and low stray inductance and resistance.

So who knows it might be a win. Please let me know what you all think.

Kind Regards,
Sparsh

[nuno]

I don't see a point going much over 16 - 18KHz (if you don't like to ear it) switching frequency, unless your motor's phase has some really low inductance. Higher switching frequency -> higher losses. Inductive switching dissipation can be estimated by F_PWM * (T_sw(ON) + T_sw(OFF)) * V_ds * I_d * 0.5

[rx8pilot]

I don't see a point going much over 16 - 18KHz (if you don't like to ear it) switching frequency, unless your motor's phase has some really low inductance. Higher switching frequency -> higher losses. Inductive switching dissipation can be estimated by F_PWM * (T_sw(ON) + T_sw(OFF)) * V_ds * I_d * 0.5

The overall size of the solution is the dominant reason. Ripple current can also be drastically reduced with higher switching speed.

It is a giant pile of parameters that have to balance to best meet the needs of the final product. Size, cost, noise, transient performance, ripple currents, thermal budget, etc,etc.... Kind of like spinning a dozen plates while walking a tightrope.

[nuno]

How will the size of the solution be reduced with increased switching frequency?

[rx8pilot]

The inductors and capacitors grow.

[nuno]

In a DC-DC converter yes, but a motor controller is a little different. The inductor has fixed size and is outside the board (the motor). The bulky capacitors can't be reduced that directly, because the power wiring doesn't change, and there's the risk of resonance. On the other hand, higher dissipation -> bulkier heatsinking.
I would say use the lowest switching frequency possible that keeps your motor current ripple at an acceptable level.

[rx8pilot]

In a DC-DC converter yes, but a motor controller is a little different. The inductor has fixed size and is outside the board (the motor). The bulky capacitors can't be reduced that directly, because the power wiring doesn't change, and there's the risk of resonance. On the other hand, higher dissipation -> bulkier heatsinking.
I would say use the lowest switching frequency possible that keeps your motor current ripple at an acceptable level.

You are right - I totally forgot this project is a motor drive. 

[T3sl4co1l]

Assuming you have short and/or shielded cables leading to the motor, of course -- otherwise with EMI filtering, it basically constitutes a rather powerful class D audio amplifier, and you might as well push up the frequency!

Tim

[max_torque]

For motor control, you'll be running your d/q axis current control function locked into the fundamental PWM loop.  Because you'll want to change the phase voltage modulation at every opportunity.  If you run too fast, you won't be able to fit your control code into the time available and hence you now will have several PWM cycles with the same modulation, which defeats a lot of the purpose of running faster!

For pretty much all practical motors, using centre aligned pwm, with it's intrinsic frequency doubling, means that the ripple currents will already be very small at most sensible fundamental pwm frequencies (8 to 20 khz).  Of course, if you are driving some sort of ultra-trick, ultra low inductance high speed eMachine then sure, start ramping the frequencies up.

And of course, honorable mention here to EMC compliance?  IS it required, to what level? The faster you switch, the bigger head-ache it becomes.  Or if you switch slowly, then your deadtime becomes a larger proportion of your modulation index.