Preamp for Analog Discovery 2

[thanasisk]

If you need the channel to be fully differential, you can always just duplicate the circuit - you add some variability to the system, but given the bandwidth you can probably just make sure your gain resistors are tight tolerance and your opamps have low input offset voltage and you'll probably be fine so long as the two are colocated on a board or in the same enclosure.  If you need a differential input, you can duplicate the first low gain stage and then send each into a secondary amp to move to single ended and do your final gain stage with that.  If you want differential input into the analog discovery for whatever reason but are fine with single ended inputs, you can just have an inverting follower amp after the output stage to get a mirrored signal (though I don't think it's going to promote signal integrity unless you need big time EMF rejection between the output of the preamp and the input of the analog discovery).

Personally, I don't know why you'd need a differential signal for this application, but perhaps your situation can demand it.  I would certainly prefer two distinct channels to play with and just go into a differential mode in software, and at these frequencies and signal levels, unless your area is very noisy in terms of EMF (and if it is, you probably need to shield the analog discovery too), single ended should be able to give you plenty of signal integrity and responsiveness within the bandwidth you're looking at and will be cheaper to implement (more choices of parts, cheaper options, fewer parts).

A roll-my-own instrumentation amp with single ended ouput would make much sense. I will definitely try this,also with david's pointers.  I read somewhere (cant remember exactly where) that common mode rejection performance is bounded by the tolerance of the matched resistors and even selecting 0.1% resistors does not work wonders in that aspect. But other design aspects would be more important anyway.

[thanasisk]

What do you mean by oscilloscope "hook"? (obviously it is not the hook connectors for the probes..)

Getting Rid of Hook: The Hidden PC-Board Capacitance

You probably will not have a problem but it is important where high impedance dividers are used like on the AD2.

Looks like a complex problem to address. I trust that they dimensioned  the variable capacitor sufficiently in the AD2 (5-20pf) to roughly compensate for all these effects. As a side note, AD2s come precalibrated from the factory but it could be that they would also benefit from recalibration every now and then, especially after severe environmental  changes or aging.

[thanasisk]

8

Is fast overload recovery a requirement?  Shunt feedback amplifiers have problems with this.

Fast overload recovery is not a requirement (how fast are we talking here?) But I have always been wondering whether there exists is a distortion free, noise free and "cheap" way to indicate overload  with e.g. an led or digital output..

[David Hess]

I read somewhere (cant remember exactly where) that common mode rejection performance is bounded by the tolerance of the matched resistors and even selecting 0.1% resistors does not work wonders in that aspect. But other design aspects would be more important anyway.

DC and low frequency AC common mode rejection can be handled but high frequency AC common mode rejection is a major problem which involves both amplitude and phase.  Those parts I listed do not rely on matched impedances in a feedback network so have much better high frequency AC common mode rejection than would normally be the case.  High performance differential probes use a similar design.

Getting Rid of Hook: The Hidden PC-Board Capacitance

You probably will not have a problem but it is important where high impedance dividers are used like on the AD2.

Looks like a complex problem to address. I trust that they dimensioned  the variable capacitor sufficiently in the AD2 (5-20pf) to roughly compensate for all these effects. As a side note, AD2s come precalibrated from the factory but it could be that they would also benefit from recalibration every now and then, especially after severe environmental  changes or aging.

Compensating for hook in the circuit is possible but complex and hook often changes with humidity so this is not always feasible; the better option is to use a printed circuit board substrate which does not suffer from hook.  Before good FR4 substrates became available, Tektronix used special materials like polysulfone and an unknown white plastic substrate for their high impedance attenuators.

Fast overload recovery is not a requirement (how fast are we talking here?) But I have always been wondering whether there exists is a distortion free, noise free and "cheap" way to indicate overload  with e.g. an led or digital output..

The 7A13 uses feedback from the output to the input to clamp overload improving recovery time which could certainly be detected; the feedback only occurs when overload is present.  Monitoring the common mode and differential levels at the right points is enough to detect overload.

Overload is sneaky because not only does it "wind up" the integration stage used in feedback amplifiers, it also drives transistors into saturation or cutoff resulting in temperature changes which takes microseconds to milliseconds to resolve.

[PartialDischarge]

I wonder to what extent hook is still a problem with todays pcb manufacturing methods. Maybe nothing has changed from the 70s but modern literature on this topic seems inexistent.
Also the article mentions teflon standoffs as a way to support critical components whereas they don't mention slot cuts, todays CNCs make them in any shape and width, that is my preferred method of 'isolating' high impedance points.
The article also doesn't mention how the orientation of components in the board my affect performance in some cases, specially in HV circuits, this has to do with the orientation of fibers in the PCB

[David Hess]

I wonder to what extent hook is still a problem with todays pcb manufacturing methods. Maybe nothing has changed from the 70s but modern literature on this topic seems inexistent.

Nothing has changed for FR4 substrates and hook is still a problem; it is just that most circuits are not susceptible so nobody notices.  There have been reports of high voltage differential probes drifting out of their specifications which seem suspiciously like a problem with hook.  I have noticed it in high impedance circuits which do not settle as quickly as they should or mysterious non-linearity in high resolution analog to digital converters.  I suspect the designers of multimeters run across it in their high impedance dividers leading to errors in AC measurements at different frequencies.

Where the lack of hook matters, it is necessary to either qualify board manufacturers or use substrates which are guaranteed to have a low level of hook.

Also the article mentions teflon standoffs as a way to support critical components whereas they don't mention slot cuts, todays CNCs make them in any shape and width, that is my preferred method of 'isolating' high impedance points.

Usually the concern is controlling leakage.

The article also doesn't mention how the orientation of components in the board my affect performance in some cases, specially in HV circuits, this has to do with the orientation of fibers in the PCB

I know this is important for controlling the impedance of transmission lines at high frequencies on woven substrates but why would it matter for high voltage circuits?

[PartialDischarge]

Beacuse the dielectric strength varies from one axis to another, I’m talking under many tens of kVs, with pcbs operating in transformer oil, the limiting factor becomes not creepage or clearance but the paths inside the pcb through the fiber

[CZ101]

Great thread.

I have been investigating options for a high impedance differential preamp for looking at tiny signals in the audio band. My application right now is for probing differential microphone-level signals, somewhere between 5-50 mV RMS. The output would be single ended going into a typical scope 1 meg input.

The Tek AM502 would probably be exactly what I'm looking for but I'd kind of like to make this a DIY learning project and also see if lower noise is achievable.

I've been thinking about using the LSK489 as a differential JFET input pair:

http://www.linearsystems.com/lsnews/lsk489News.html
http://www.linearsystems.com/lsdata/datasheets/LSK489_LOW_NOISE,_LOW_CAPACITANCE_MONOLITHIC_DUAL_N-CHANNEL_JFET.pdf

Here's a nice app note from Bob Cordell on the LSK489:
http://www.cordellaudio.com/JFETs/LSK489appnote.pdf

The LSK489 would be in front of perhaps the AD8429 instrumentation amp http://www.analog.com/media/en/technical-documentation/data-sheets/AD8429.pdf. 1 Mhz bandwidth would be adequate, but higher would be welcome.

I'm right now trying to figure out the gain and probe matching options to optimize for CMRR and noise. The Tektronix probes with trimmable DC resistance (and capacitance) to match pairs for CMRR like the P6023 or P6135 are nice but they are 10x and it seems like these would really only be useful for larger signals, unless I am mistaken.

Then again the differential amplifier's rti noise decreases with gain, so maybe I am looking at this the wrong way and should go for 10x probes.

There is also the option of simply using a shielded twisted pair and making a diy probe that way.

[David Hess]

Beacuse the dielectric strength varies from one axis to another, I’m talking under many tens of kVs, with pcbs operating in transformer oil, the limiting factor becomes not creepage or clearance but the paths inside the pcb through the fiber

To prevent ionization where the dielectric constant falls?  I know this is an issue with bubbles in the potting material used in high voltage assemblies.

[David Hess]

I have been investigating options for a high impedance differential preamp for looking at tiny signals in the audio band. My application right now is for probing differential microphone-level signals, somewhere between 5-50 mV RMS. The output would be single ended going into a typical scope 1 meg input.

The Tek AM502 would probably be exactly what I'm looking for but I'd kind of like to make this a DIY learning project and also see if lower noise is achievable.

I've been thinking about using the LSK489 as a differential JFET input pair

At lower frequencies shunt feedback is more acceptable so you end up with a design like the Tektronix 5A22/7A22/AM502 using a differential JFET pair with at least enough gain to overcome the noise of the following low impedance stages.

If you do not want to mess around with the LSK489 and similar which will require considerable support circuitry, then parallel low noise JFET input operational amplifiers might be an acceptable simplification.

There is something to watch out for with dual JFETs however; this does not matter at audio frequencies but at higher frequencies, electrical coupling between the monolithic JFETs can ruin common mode rejection and transient response.  Jim Williams mentioned this in his own differential probe amplifier design shown below.  This may be why Tektronix used matched pairs in the 100MHz 7A13 instead of a dual part.  The slower 1MHz 5A22/7A22/AM502 used a dual part.

The Jim Williams design shown below is one of those with horrible overload recovery because of the slow DC common mode stabilizer loop.  It also is not optimized for low noise; there is no voltage gain in the JFET stage like with the 5A22/7A22/AM502.  Neither of these things mattered for his application.  Also note that he used one of those magical difference amplifiers on my list for differential to single ended conversion.

The LSK489 would be in front of perhaps the AD8429 instrumentation amp http://www.analog.com/media/en/technical-documentation/data-sheets/AD8429.pdf. 1 Mhz bandwidth would be adequate, but higher would be welcome.

Notice that there is a tradeoff with input noise, gain, and bandwidth for the AD8429.  If the JFET preamplifier has enough gain to overwhelm the noise, then the AD8429 can be operated at a lower gain resulting in increased bandwidth.

I'm right now trying to figure out the gain and probe matching options to optimize for CMRR and noise. The Tektronix probes with trimmable DC resistance (and capacitance) to match pairs for CMRR like the P6023 or P6135 are nice but they are 10x and it seems like these would really only be useful for larger signals, unless I am mistaken.

Then again the differential amplifier's rti noise decreases with gain, so maybe I am looking at this the wrong way and should go for 10x probes.

10x probes have two problems; the attenuation raises the input noise by the same amount (and more due to the 9 megohm series resistor but this is reduced at higher frequencies by the parallel capacitance) and the mismatch in attenuation lowers the common mode rejection a lot unless special matched probes are used.  The economical way to get around the matching issue is simply to add DC and AC trims to the 1 megohm shunt network at the preamplifier use a dedicated set of 10x probes.  The trimmable dual probes which Tektronix made did this with a network at the compensation box but had to pay for it by lowering the input series resistance from 9 megohms to a lower value.

There is also the option of simply using a shielded twisted pair and making a diy probe that way.

Or use 1x probes which will not degrade the input noise or common mode rejection.

Often you need some attenuation to get enough common mode input voltage range unless AC coupling is acceptable or your differential amplifier has a lot of input common mode range.  Unlike most oscilloscope inputs and bare active probes, the 5A22/7A22/AM502 and 7A13 have an input common mode range of +/-10 volts and the design we have been discussing with thanasisk also could have that much range just by using the right JFET operational amplifier and the AD830/AD8129/AD8130.

[Marco]

The LSK489 would be in front of perhaps the AD8429 instrumentation amp

Using the JFET only as an impedance transformer seems like a shame. There are circuits which preserve the noise level of the JFET.

[PartialDischarge]

Beacuse the dielectric strength varies from one axis to another, I’m talking under many tens of kVs, with pcbs operating in transformer oil, the limiting factor becomes not creepage or clearance but the paths inside the pcb through the fiber

To prevent ionization where the dielectric constant falls?  I know this is an issue with bubbles in the potting material used in high voltage assemblies.

Ionization and flashover. Sometimes ionization is not that much of a problem, like in equipments with low duty cycle, X-ray machines for human use generate up to 150kV and the duty is very low even in continuous fluoroscopy. However a flashover will make noise and sometimes render the equipment useless, maybe not by a technical fault per se, but as a reminder that high voltage has found its way around and will find it again.

Bubbles create a different problem, partial discharges. In oil there are no partial discharges since the electric field will shape a bubble, a droplet of water or a hair the way it wants to, to create flashover. In solid materials, like the epoxies used in instrument transformers, bushings or busbar insulators, a bubble or a small pebble inside represent a change in relative permittivities, which implies a redistribution of the field lines in that area. It's like having a capacitive divider inside the insulator. At some voltage, the inception voltage, the air in the bubble breaks down creating UV and ozone. With enough time the insulator develops cracks and fails. High voltage is like Tim Robbins in the Shawshank Redemption, it takes time but it will eventually escape.

High voltage 101 is thinking how the field and equipotential lines will behave. And they do according to the materials that they encounter. Sometimes a common mistake in high voltage systems is thinking that a thick piece of insulating material in between two air-insulated high voltage conductors improves reliability and that could be false, since insulating material with higher permittivity will drive away field lines to the outside, where now they become more condensed and breakdown could occur. Or place a low permittivity material inside a high valued one, and it will act as the small capacitor in a capacitive divider.

[David Hess]

The LSK489 would be in front of perhaps the AD8429 instrumentation amp

Using the JFET only as an impedance transformer seems like a shame. There are circuits which preserve the noise level of the JFET.

Both of those examples work the same way as the 5A22/7A22/AM502.  That is an awful lot of extra complexity for a 3dB noise improvement over JFET source followers driving a AD8429 configured for high gain.

I would like to see what kind of bandwidth that design driving a much higher noise AD8129  could produce.  I suspect it leads to all kinds of problems making JFET source followers desirable despite higher noise.

[thanasisk]

Have been quietly working on simulating a preamp based on LT1102 to have as a comparison reference against further designs that we have been discussing here,  however I do not seem to be able to achieve bandwidths of more than 50/550kHz (G of 100/10) despite the datasheet indication of approx 5 times that BW. Also the input and output noise is sky high.  Makes me wonder whether the pspice model from AD is ok or whether I am doing an obvious mistake in the design. Will post here my schematic hopefully late in the evening if time allows..

[David Hess]

When you post the schematic, also list the test conditions of the LT1102 circuit.

SPICE models do not always model every aspect of a part.  The better ones include a list of what is modeled.

[thanasisk]

Just throwing this in the pile, maybe it'll be useful to you at some point.

If you want to go all the way down to DC and up to say 10 or 20 Hz,  you can use a split path amp. You can have you ac-coupled amp,  say 10Hz to 1MHz  then,  also have something like a LTC1050  handling the DC part,  in parallel. Identical gain,  of course.

There's something  regarding this in App Note 106 fron  Linear.

There is also Application Note 47 from Linear:

http://www.analog.com/media/en/technical-documentation/application-notes/an47fa.pdf

They use a split path (Fig 74, 76 for gains of 10 and 1000 respectively). Interestingly an LT1102 handles the DC path, and a video difference high input impedance amplifier the AC path.

[thanasisk]

When you post the schematic, also list the test conditions of the LT1102 circuit.

SPICE models do not always model every aspect of a part.  The better ones include a list of what is modeled.

Here is the spice model. Not completely sure what is modelled and what is not:

http://www.analog.com/media/en/simulation-models/spice-models/LT1102.txt

I am attaching the schematic along with bode and noise plots (for several values of source impedance). I also attach the tina-ti/spice "analysis parameters".

I hope I am not doing an obvious stupid mistake with the schematic /design. G=20 as expected but BW much smaller than the data sheet. And noise is enormous.

[thanasisk]

Any ideas or feedback about the circuit? 

[Marco]

Properly DC bias the inputs, not through the diodes.

[David Hess]

Properly DC bias the inputs, not through the diodes.

Yep, I am not sure what SPICE does with unbiased JFET inputs but it probably is not good; either move the 1 megohm resistors to the amplifier side of the input capacitors for testing purposes or do what an oscilloscope input does and place like 100 kilohm resistors across the input capacitors.

1N4004 diodes are not suitable due to excessive leakage and capacitance.  People seem to like the BAV199 dual low leakage diode (3pA 2pF) but I like 2N3904 collector-base junctions.  The input capacitance of the protection diodes can be halved by using pairs in series for each one.

Note that the slow reverse recovery of the low leakage input protection diodes screws up overload recovery time.  Switching diodes like the 1N4148 are much faster but the gold doping which makes them fast also makes them have too much leakage for an application like this.  The base-emitter junction of a 2N3904 is both low leakage and fast but has a very limited reverse breakdown voltage.  They used to make 15 volt fast low leakage low capacitance diodes but there are other ways to do it if fast recovery of the input protection circuits is a requirement.

[thanasisk]

Well, Marco & David,  I placed the bias resistors after the input caps (see attached schematic) or even at the + and - inputs of LT1102 ; But this had no effect on the simulation results.

David I had in mind  the -GP version 1N4004GP which is Glass Passivated with 8pF junction capacitance, 5uA reverse current (at 100deg celcius at 400V peak reverse voltage)/(or just 20-50nA at 25deg celcius). Reverse recovery is 2usec though.

Why do you consider the 1N4148 as unsuitable? The worst case leakage current of 50uA is specified at 150 degrees celcius  BUT  at a more normal 25degrees it is in the region of 20-100nA depending on the (reverse) voltage.

Or do you follow worst case specs because you can expect a sudden junction temperature increase during overvoltage events?

As a side note, no reasonably priced through-hole part can come close to the specs the BAV199 that you mentioned.

[thanasisk]

And here is another take with an INA111. With similar layout as the previous posts for the LT1102. But it works this time! I will recheck the importing of the LT1102 model..

With the INA111 a gain of 2 brings the noise at the output above the noise floor of the AD2.

So I can now look for a suitable opamp with a gain of 10 and 100 to see what I can get. 

[David Hess]

David I had in mind  the -GP version 1N4004GP which is Glass Passivated with 8pF junction capacitance, 5uA reverse current (at 100deg celcius at 400V peak reverse voltage)/(or just 20-50nA at 25deg celcius). Reverse recovery is 2usec though.

But maximum reverse current at room temperature is 5 microamps and reverse current is only weakly related to reverse voltage.  The 8 picofarads of capacitance gets doubled with two diodes resulting in a total input capacitance high enough to present problems with probe compensation and bandwidth.

Why do you consider the 1N4148 as unsuitable? The worst case leakage current of 50uA is specified at 150 degrees celcius  BUT  at a more normal 25degrees it is in the region of 20-100nA depending on the (reverse) voltage.

20 nanoamps into 1 megohm is 20 millivolts and this leakage varies directly with temperature.

Worst case high temperature leakage current into the LT1102 inputs is about 2 nanoamps and leakage current at room temperature is more like 100 picoamps.

Or do you follow worst case specs because you can expect a sudden junction temperature increase during overvoltage events?

Leakage from junction temperature increases due to overload is actually a consideration for fast overload recovery.

It is worth mentioning where these relatively high absolute maximum leakage specifications come from.  It is expensive to test for low leakage (tester time is charged per second) so a specification like 5uA or 20nA represents what the tester itself was capable of within the time allowed for testing.  Where the LT1102 datasheet says +/-60pA maximum at 25C, +/-400pA maximum at 70C, and 15nA maximum at 125C, they are not kidding even though the typical specifications might be 10 times better.  2N3904 base-collector junctions are specified to be 50nA at 25C because of the test itself but are typically more like 10pA and better.

As a side note, no reasonably priced through-hole part can come close to the specs the BAV199 that you mentioned.

Dedicated low leakage diodes were never commonly available and technically the BAV199 does not count as low leakage either because it is only tested to be less than 5 nanoamps.

Manufacturers have used small signal transistor base-collector or base-emitter junctions as low leakage diodes for decades.  Like the BAV199, they are not tested for low leakage either but demand was never high enough to manufacture a tested low leakage part at an economical price.  The user has to qualify or test them themselves which is not difficult.

If you want an inexpensive already tested low leakage diode, then a 2N4117/2N4118/2N4119 low input bias current JFET (10pA at 25C maximum and 1pA typical) is the most economical choice and a lot of manufacturers use low leakage JFETs for exactly this purpose; just tie the drain and source together.

And here is another take with an INA111. With similar layout as the previous posts for the LT1102. But it works this time! I will recheck the importing of the LT1102 model..

That is weird; I was going to suggest that the SPICE was modeling the conductance of the 1N4004GP diode attenuating the input but obviously there is something wrong with the LT1102 model.  At zero volts, diode conductance is about 26 mhos/amp. (1)

(1) NIST's Guide for the Use of the International System of Units (SI) refers to the mho as an "unaccepted special name for an SI unit", and indicates that it should be strictly avoided. - Fuck you NIST.  I no longer accept any of your advice since you approved deliberately compromised NSA algorithms and standards for cryptography.  (2) You are not to be trusted with anything.  You are as bad as the FDA.  Die in a fire.

(2) Yea, this is just an excuse.  I would use mho in place of siemen anyway.  But NIST really is no longer to be trusted at least with anything having to do with cryptography and computer security.  They can still die in a fire.

[thanasisk]

David I had in mind  the -GP version 1N4004GP which is Glass Passivated with 8pF junction capacitance, 5uA reverse current (at 100deg celcius at 400V peak reverse voltage)/(or just 20-50nA at 25deg celcius). Reverse recovery is 2usec though.

But maximum reverse current at room temperature is 5 microamps and reverse current is only weakly related to reverse voltage.  The 8 picofarads of capacitance gets doubled with two diodes resulting in a total input capacitance high enough to present problems with probe compensation and bandwidth.

Ah yes, I see that the curves in the data sheet are "TYPICAL CHARACTERISTICS".

Why do you consider the 1N4148 as unsuitable? The worst case leakage current of 50uA is specified at 150 degrees celcius  BUT  at a more normal 25degrees it is in the region of 20-100nA depending on the (reverse) voltage.

20 nanoamps into 1 megohm is 20 millivolts and this leakage varies directly with temperature.

Worst case high temperature leakage current into the LT1102 inputs is about 2 nanoamps and leakage current at room temperature is more like 100 picoamps.

Or do you follow worst case specs because you can expect a sudden junction temperature increase during overvoltage events?

Leakage from junction temperature increases due to overload is actually a consideration for fast overload recovery.

It is worth mentioning where these relatively high absolute maximum leakage specifications come from.  It is expensive to test for low leakage (tester time is charged per second) so a specification like 5uA or 20nA represents what the tester itself was capable of within the time allowed for testing.  Where the LT1102 datasheet says +/-60pA maximum at 25C, +/-400pA maximum at 70C, and 15nA maximum at 125C, they are not kidding even though the typical specifications might be 10 times better.  2N3904 base-collector junctions are specified to be 50nA at 25C because of the test itself but are typically more like 10pA and better.

As a side note, no reasonably priced through-hole part can come close to the specs the BAV199 that you mentioned.

Dedicated low leakage diodes were never commonly available and technically the BAV199 does not count as low leakage either because it is only tested to be less than 5 nanoamps.

Manufacturers have used small signal transistor base-collector or base-emitter junctions as low leakage diodes for decades.  Like the BAV199, they are not tested for low leakage either but demand was never high enough to manufacture a tested low leakage part at an economical price.  The user has to qualify or test them themselves which is not difficult.

If you want an inexpensive already tested low leakage diode, then a 2N4117/2N4118/2N4119 low input bias current JFET (10pA at 25C maximum and 1pA typical) is the most economical choice and a lot of manufacturers use low leakage JFETs for exactly this purpose; just tie the drain and source together.

This is all very useful information, thank you!

All this discussion and using a jfet or transistor as diode replacement reminds me of the "biased diode clipping circuit" and the transistor clippers (e.g. BC456 and BC556 NPN/PNP transistors for diode replacement) analyzed in Chapter 24 of Small Signal Audio Design by Douglas Self. There he also mentions that circuits like that have a very high output impedance and thus he always includes an followup op-amp buffer. I wonder whether such a concern is relevant to our case here.

I will certainly try to test the JFET solution though, any pointers to relevant schematics of manufacturers?

And here is another take with an INA111. With similar layout as the previous posts for the LT1102. But it works this time! I will recheck the importing of the LT1102 model..

That is weird; I was going to suggest that the SPICE was modeling the conductance of the 1N4004GP diode attenuating the input but obviously there is something wrong with the LT1102 model.  At zero volts, diode conductance is about 26 mhos/amp. (1)

(1) NIST's Guide for the Use of the International System of Units (SI) refers to the mho as an "unaccepted special name for an SI unit", and indicates that it should be strictly avoided. - Fuck you NIST.  I no longer accept any of your advice since you approved deliberately compromised NSA algorithms and standards for cryptography.  (2) You are not to be trusted with anything.  You are as bad as the FDA.  Die in a fire.

(2) Yea, this is just an excuse.  I would use mho in place of siemen anyway.  But NIST really is no longer to be trusted at least with anything having to do with cryptography and computer security.  They can still die in a fire.

Well as long as you pass on unambiguous info, any term should be fine

I further tested an opamp post-amp configuration and will post my findings here soon. I got stuck a bit because of the compensation and because I also want to simulate for adequate THD+N performance..

[David Hess]

If you want an inexpensive already tested low leakage diode, then a 2N4117/2N4118/2N4119 low input bias current JFET (10pA at 25C maximum and 1pA typical) is the most economical choice and a lot of manufacturers use low leakage JFETs for exactly this purpose; just tie the drain and source together.

All this discussion and using a jfet or transistor as diode replacement reminds me of the "biased diode clipping circuit" and the transistor clippers (e.g. BC456 and BC556 NPN/PNP transistors for diode replacement) analyzed in Chapter 24 of Small Signal Audio Design by Douglas Self. There he also mentions that circuits like that have a very high output impedance and thus he always includes an followup op-amp buffer. I wonder whether such a concern is relevant to our case here.

The use of the transistors instead of diodes in Audio Design (great book, recommended) was because of higher conductance producing a slightly sharper knee in the clamp circuit.  Audio circuits have much different requirements than oscilloscope inputs because they can tolerate much less distortion.

Diode conductance (26 mhos/amp at zero volts) is a real thing and in high impedance circuits it can cause real problems if low leakage diodes are not used.  For oscilloscopes, up to a certain point capacitance is more important given the frequency requirements.  You *could* get away with gold doped high leakage 1N4148s but their leakage would cause considerable DC drift over temperature at 1mV/div in a 1 megohm high input impedance amplifier or buffer.

Oscilloscope front ends are not particularly linear and suffer from considerable distortion in the quest for high bandwidth and good transient response.  A front end intended for low distortion measurements not far from the audio band requires a different circuit topology including different input protection circuits.  For low distortion where input protection was still required, I would look to bootstrapping the protection circuits themselves so the clamp voltages follow the input voltage up to the clamp limits.  This is common in high performance multimeters and even some oscilloscopes do this.

I will certainly try to test the JFET solution though, any pointers to relevant schematics of manufacturers?

I am not sure what you are asking.  The 2N4117/2N4118/2N4119 JFETs (and the A versions) have been the lowest cost guaranteed low leak current diodes for a long time.  Other JFETs will work as well but most are not specifically tested for low leakage unless they are intended for applications which require low leakage.