EEVblog Electronics Community Forum
EEVblog => EEVblog Specific => Topic started by: EEVblog on July 06, 2020, 11:10:29 pm
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Is there a better chopper/auto-zero amplifier than the MAX4239 used in the uCurrent?
A look at the exciting world of parametric searching for components. Follow along as Dave looks for a new state-of-the-art opamp. Success not guaranteed!
https://www.youtube.com/watch?v=y5qjGPZ5VhQ (https://www.youtube.com/watch?v=y5qjGPZ5VhQ)
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I agree with you that it's a lot of fun to research new parts like this.
I think current-sens amplifiers might be the wrong category, because you are not measure currents and you are not interested in low input bias currents etc.
In your application you are measuring low voltages, hence try looking into (and please make a video) on nano-voltmeters! Keithley have some nice documents on this, and you should be able to find circuit diagrams of some of the older models.
On a related topic, I recently developed a fC charge amplifier for DC to a few Hz. The best I could find for such applications was the impressive LMP7721 with 3 fA bias current and 26 uV offset voltage (very low for such an amplifier). It is easy to forget that burden/offset voltage is very important for low current measurements as any offset allows for voltage differences which causes leakages, even with guarding.
Edit:
Of course nano-voltmeter type circuits are usually only for low frequency so perhaps not applicable, but interesting nevertheless.
Another option would be transfer impedance amplifier circuit (at least for 10 mA and below) = Close to zero burden voltage at any current and could easily work down to pA currents (with reduced bandwidth).
I found this article from TI where they design a 1 mV/nA amplifier with 100 kHz bandwidth powered from +-2.5 V rails (very close to the lowest range on your uCurrent):
https://e2e.ti.com/cfs-file/__key/telligent-evolution-components-attachments/01-930-00-00-00-66-60-61/Op-Amp-Bandwidth-for-Transimpedance-Amplifiers.pdf
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Check this low cost part,TP5552 from 3PEAK, it's around 5uV but it's under 0.6USD for 10 units.
https://lcsc.com/product-detail/Others_3PEAK-TP5552-VR_C248604.html
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Check this low cost part,TP5552 from 3PEAK, it's around 5uV but it's under 0.6USD for 10 units.
https://lcsc.com/product-detail/Others_3PEAK-TP5552-VR_C248604.html (https://lcsc.com/product-detail/Others_3PEAK-TP5552-VR_C248604.html)
The offset voltage distribution graph has a weird horizontal scale, but otherwise suggests the part is much better than the max of 5uV.
I saw some other interesting parts by 3peak on LCSC once. Their datasheets seem easy to read (I think they're copying some of the better layout styles), but I don't know anything more about them.
EDIT: Main website: http://www.3peakic.com.cn/En (http://www.3peakic.com.cn/En)
EDIT2: Their low drift (http://www.3peakic.com.cn/En/product/productes/catid/124.html) parts seem to have low Voffsets, but their low noise (http://www.3peakic.com.cn/En/product/productes/catid/125.html) parts don't. Categories are probably arbitrary anyway.
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The AEC-Q 100, 101, 200 standards describe, how electronic components (IC, discrete and passive components) have to be qualified /tested by the manufacturers before they are allowed to sell their parts to the Automotive Electronics industry, like Continental, Bosch, Valeo, Visteon, and so on. The tests include environmental, reliability and electrical tests.
These are the de facto automotive industry standards, i.e. very important for us. Go to www.aecouncil.com (http://www.aecouncil.com) for details, and check the 3 base documents.
Specially for Automotive, manufacturers specify and test for -40.. +85°C for Interior, and -40 .. +125°C for Power Train, Body Control and Safety applications, which you already found in all of the chopper specifications.
Use of the standard Industrial range of 0..+70°C is mostly not allowed in Automotive.
This whole subject about electronic components in Automotive electronics, tests, reliability, Quality Systems, changes and obsolescence, and so on might give enough content for an interesting EEVblog.
Frank
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If bandwidth is important, one could change the circuit from the 2 x10 stages to one compound amplifier with a 2nd non AZ inside the loop of the AZ OP. So the other OP could be some 20-50 MHz GBW and do most of the gain.
Another advantage would be saving 2 precision resistors for the gain.
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I appreciate a large part of the video is to show ways to go about finding suitable parts and understanding datasheets and the actual devices chosen and specs are less important. However I think your concentration on offset voltage is a case of not seeing the woods for all the trees. The MAX4239 seems to be a rather poor choice for the ucurrent - it's far too noisy at 30nV/sqrt(Hz). Given the ucurrent's specified 300khz bandwidth, input noise is 108uVpp! (This is inline with the 10mV output noise reported in these Forums). Offsets below, say 10uV, are lost in the noise.
The ADA4522, ADA4528 and OPA189 are better at around 5 to 10nV/rt(Hz) but the noise varies with frequency - the 4528 having a large noise peak at 200kHz. The new ADA4523 is better still with 0-300kHz integrated noise of 20uVpp and an outstanding 88nVpp 0.1 to 10Hz. You missed this one because the maximum offset is 4uV, but if you look at the distribution graph, all 160 samples were within +/- 0.5uV offset at a supply voltage of +/- 2.5V. The 4.5mA supply current is perhaps too high for a coin cell supply (but the second amp can be a lower power device as it's noise and offset contributions are only 1/10 of the first). The ADA4523 is cheap at $0.85 @1k.
None of these alternatives were available at the time the current was designed, so the MAX4239 may well have been the best choice for the circuit architecture chosen, but a better option would probably have been to use a hybrid amp using a low noise fet amp with a zero drift/chopper for DC stabilization.
For DC applications, such as using the current with a slow DMM, most of the noise is filtered out but again the MAX4239 has a LF noise spec of 1.5uV. Added to the 2.5uV max offset spec means that lower noise, but higher offset parts may be as good if not better.
Another problem I have with the video was the discussion of the distribution charts. These are typical characteristics showing parts from early production. I don't know what real value, if any, these provide. They aren't guaranteed and any tweeks or changes to the production process can render them useless. You would be foolish to rely on them except perhaps for choosing a part that you will routinely screen as part of your manufacturing process. Even then the supplier could change the process such that the yields of parts close to the typical values you were relying on become so low as to render your product unviable.
Data sheets often have serious inconsistencies between typical values listed in the specifications and those shown in the typical graphs and some manufacturers have a reputation for supplying parts that almost never come close to their typical values.
Finally, the GBW isn't the whole story - the ADA4522 has a GBW of 2.7MHz but a closed loop bandwidth at a gain of 1 of 6.5MHz. But why does the ucurrent only have 300kHz bandwidth from the two x10 6.5MHz GBW amps?
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I appreciate a large part of the video is to show ways to go about finding suitable parts and understanding datasheets and the actual devices chosen and specs are less important.
Like Dave apparently, I really enjoy doing parts selection and comparison and how it affects design. I often end up making big spreadsheets comparing parts with datasheet specifications, calculated specifications, and measured specifications.
However I think your concentration on offset voltage is a case of not seeing the woods for all the trees. The MAX4239 seems to be a rather poor choice for the ucurrent - it's far too noisy at 30nV/sqrt(Hz). Given the ucurrent's specified 300khz bandwidth, input noise is 108uVpp! (This is inline with the 10mV output noise reported in these Forums). Offsets below, say 10uV, are lost in the noise.
I was going to say the same thing. There is a place for a chopper stabilized amplifier in this application, but the design has some contradictions. See below.
None of these alternatives were available at the time the current was designed, so the MAX4239 may well have been the best choice for the circuit architecture chosen, but a better option would probably have been to use a hybrid amp using a low noise fet amp with a zero drift/chopper for DC stabilization.
That is exactly what should be done unless the bandwidth is significantly limited, which contradicts using two operational amplifiers in series to boost bandwidth. That is the contradiction I mentioned above; the high broadband noise from the chopper stabilized amplifier limits the usefulness of higher bandwidth. Using a chopper stabilized amplifier alone is less expensive though.
If it were not for the 10 kilohm source resistance at the highest sensitivity, then a low noise bipolar part would be suitable but in this case, a low noise JFET part, or discrete JFET or MOSFET input stage will allow for almost 2 orders of magnitude better sensitivity while maintaining the same drift by using the chopper stabilized operational amplifier to remove drift and flicker noise.
For DC applications, such as using the current with a slow DMM, most of the noise is filtered out but again the MAX4239 has a LF noise spec of 1.5uV. Added to the 2.5uV max offset spec means that lower noise, but higher offset parts may be as good if not better.
I laughed at that "low noise" claim on the datasheet. Chopper stabilized amplifiers are only lower noise than the alternatives at much lower frequencies. The datasheet for a chopper stabilized amplifier lists noise from 1 to 10 Hz only for comparison purposes for comparison with other chopper stabilized amplifiers. What matters in practical applications is noise from DC to 1 Hz so if this is not listed, beware.
Data sheets often have serious inconsistencies between typical values listed in the specifications and those shown in the typical graphs and some manufacturers have a reputation for supplying parts that almost never come close to their typical values.
Datasheets sometimes outright lie, and often mislead as with the "low noise" part mentioned above, or listing offset in microvolts rather than nanovolts as Dave noticed. Texas Instruments has a history going back decades of doing that sort of thing but they are not the only one.
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The high bandwidth of the max4239 is of limited use because of the high noise - though there are more noisy ones. At higher frequencies the max4239 is not that bad for an AZ OP with supposedly some 30 nV/sqrt(Hz). Still a scope input amplifier may be lower noise for the higher frequencies, or at least comparable.
There is a difference between classical auto-zero OPs (e.g. the max4239, LTC1052, TLC2652) and real chopper stabilized OPs (e.g. OPA189). They have a different noise spectrum. For a high bandwidth design the copper signals in a copper stabilized OP can be a real problem.
A separate low noise amplifier for the high frequency part would indeed make sense. It would make the circuit more complicated and may need some trimming (e.g. to get a smooth transition).
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The high bandwidth of the max4239 is of limited use because of the high noise - though there are more noisy ones. At higher frequencies the max4239 is not that bad for an AZ OP with supposedly some 30 nV/sqrt(Hz). Still a scope input amplifier may be lower noise for the higher frequencies, or at least comparable.
May? MAY? A scope input amplifier better have a lot lower noise, which is why I often laugh when marketing advertises a modern oscilloscope as being "low noise".
The ones I am familiar with are "chopper stabilized". "Chopper amplifiers" are a completely different thing. There are, or were, some other input offset voltage correcting technologies which had no switching noise at all. I am not sure about "automatic zero".
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How much bandwidth does the uCurrent actually need?
Its response seems well behaved and reasonable for its typical use cases.
[attachimg=1]
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How much bandwidth does the uCurrent actually need?
It needs to be fast enough for a multimeter's AC input ranges, which themselves are not all that flat above low frequencies.
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How much bandwidth does the uCurrent actually need?
It needs to be fast enough for a multimeter's AC input ranges, which themselves are not all that flat above low frequencies.
I guess another use case is hooking the uCurrent up to a scope, and watching the current consumption unfold on the screen... but 300KHz seems more than enough for that purpose too, in practice most devices have at least one bypass capacitor somewhere on their rail(s) that limits current pulse bandwidth...
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Good comment from youtube:
John Wettroth
1 day ago
Hi Dave, I retired as the managing director of product definition for standard products at Maxim a few years ago. My group and I defined thousands of products like this for over 20 years and I had a hand in the 4239 parts as well as many, many others. There were a couple issues that came up in this video that I can shed a bit of light on from the chip company side. Although Philbrick made precsion amps in 50's with tubes, Intersil , the spiritual father of Maxim invented this category in CMOS chopper amps in the 70's with the ICL7652. Maxim's second sourced these parts and made improved second sources early in its life. Here are a few points that may give you a better handle on how chip companies approach high performance analog. Because of the way that these parts work, the nominal offset is zero. What gets in the way of this is real third order effect like thermocouple action in the leads and package stress. Maxim is somewhat unique in that it can and does do post package trim (zener zapping) vs. just laser trim a the wafer level. Parts that are perfect at wafer end up being a couple of uV post package due to stresses. The other thing about very high spec parts like this (100 nV) is testing. In order to get good yields, test max limits are generally set higher, we called this the threshold of pain. The typicial values and histograms are there to give you a feel for what you're really going to get. In difficult to test specs, you're balancing how many seconds you're spending on a million dollar tester- the cost get significant. Some specs llike leakage currents in analog switches will often have a max spec of 10 uA though will generally be in the femto amp range. We would release a different external part number that guaranteed a spec like this. You can get an idea of the real spread by asking a vendor to make you 10,000 (a common minimum) with some spec tightly tested and they don't balk, it means the parts will yield if the customer is willing to pay the delta in test cost (plus NRE).. Maxim routinely does this for big customers with precision requirements like a test equipment company. The histograms in the data sheet is taken from the first few wafer lots as the test guys and the design guys dial things in- this process is called correlation. On a little amplifier parts on big wafers, there be 5000 die per wafer so you can generate a lot of test data pretty quickly. The voltage range/dynamic range issue is driven by modern processes. Maxim has a lot of boutique processes but most analog part in the market are made on kind of vanilla cmos processes which are generally digital and low voltage. These are generally 100-200 micron processes. If you see cmos parts with +-5V or +-15v supplies, they are fabbed on old 2 micron plus processes or a specialty analogy process. Maxim, TI and Analog and a very few others keep these old processes running for precision analog. The economics of doing 130 nm analog on 300 mm wafers makes parts really cheap even though it takes some real design chops to do it. Your comments about automotive are somewhat on target, they can be robust but mainly this designation has to do to with consistency of supply, change notices and paper. Sometimes specs are relaxed to improve yields also. Automotive guys can work with anything but they don't like surprises. Hope this helps a little.
So explains why other companies aren't near these specs, who knows maybe they even have patents.
One possible alternative, that would be relevant for hobbyist audience: test in production that the Vos is within "tweaking range" then have a note for the user to tweak it themselves on arrival.
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Wow, thankyou for finding that. Re-quoted with line-breaks:
John Wettroth
Hi Dave,
I retired as the managing director of product definition for standard products at Maxim a few years ago. My group and I defined thousands of products like this for over 20 years and I had a hand in the 4239 parts as well as many, many others. There were a couple issues that came up in this video that I can shed a bit of light on from the chip company side.
Although Philbrick made precsion amps in 50's with tubes, Intersil , the spiritual father of Maxim invented this category in CMOS chopper amps in the 70's with the ICL7652. Maxim's second sourced these parts and made improved second sources early in its life.
Here are a few points that may give you a better handle on how chip companies approach high performance analog.
Because of the way that these parts work, the nominal offset is zero. What gets in the way of this is real third order effect like thermocouple action in the leads and package stress. Maxim is somewhat unique in that it can and does do post package trim (zener zapping) vs. just laser trim a the wafer level. Parts that are perfect at wafer end up being a couple of uV post package due to stresses.
The other thing about very high spec parts like this (100 nV) is testing. In order to get good yields, test max limits are generally set higher, we called this the threshold of pain. The typicial values and histograms are there to give you a feel for what you're really going to get. In difficult to test specs, you're balancing how many seconds you're spending on a million dollar tester- the cost get significant. Some specs llike leakage currents in analog switches will often have a max spec of 10 uA though will generally be in the femto amp range. We would release a different external part number that guaranteed a spec like this.
You can get an idea of the real spread by asking a vendor to make you 10,000 (a common minimum) with some spec tightly tested and they don't balk, it means the parts will yield if the customer is willing to pay the delta in test cost (plus NRE).. Maxim routinely does this for big customers with precision requirements like a test equipment company.
The histograms in the data sheet is taken from the first few wafer lots as the test guys and the design guys dial things in- this process is called correlation. On a little amplifier parts on big wafers, there be 5000 die per wafer so you can generate a lot of test data pretty quickly.
The voltage range/dynamic range issue is driven by modern processes. Maxim has a lot of boutique processes but most analog part in the market are made on kind of vanilla cmos processes which are generally digital and low voltage. These are generally 100-200 micron processes. If you see cmos parts with +-5V or +-15v supplies, they are fabbed on old 2 micron plus processes or a specialty analogy process. Maxim, TI and Analog and a very few others keep these old processes running for precision analog. The economics of doing 130 nm analog on 300 mm wafers makes parts really cheap even though it takes some real design chops to do it.
Your comments about automotive are somewhat on target, they can be robust but mainly this designation has to do to with consistency of supply, change notices and paper. Sometimes specs are relaxed to improve yields also. Automotive guys can work with anything but they don't like surprises.
Hope this helps a little.
So the offsets mostly come from package stress and thermocouple action in the leads. I wonder how the stress contributes -- changes in the thermocouple potentials or unbalancing resistances further into the wafer?
It's also really interesting to read about the voltage limits. This explains why many low-power opamps don't go above 5.5-6V, even though it would be really useful if they could go just a bit above 4 AA batteries (6.4V).
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So the offsets mostly come from package stress and thermocouple action in the leads. I wonder how the stress contributes -- changes in the thermocouple potentials or unbalancing resistances further into the wafer?
I almost discussed this in my post above but decided it was too far from the topic.
Precision analog ICs use a special encapsulation and packaging to avoid stress on the IC die which would ruin precision. (1) With this and good design, offset voltages below 25 microvolts and offset voltage drifts below 100nV/C are feasible even on a non-chopper bipolar parts. Chopper stabilized amplifiers do slightly better than this and have the additional advantage of lower, and flat, flicker noise which tends to be an intractable problem.
But in both cases, thermocouple effects are a larger source of error, even for precision bipolar parts. So special attention has to be paid to layout and circuit design to take advantage of the potential precision available. Linear Technology application note 9 (https://www.analog.com/media/en/technical-documentation/application-notes/an09f.pdf) discusses this and shows measurements of exactly how serious this issue is.
(1) I heard an interview with ... I think it was Bob Dobkin of National Semiconductor ... who told a story about someone in management complaining about the cost of the special encapsulation which analog ICs required and asking why they could not use the same packaging as digital ICs. Some time after that, someone in management moved the analog IC packaging to the digital packaging plant and a couple months later, the engineers found out what had happened when they tracked down why the yields for analog ICs had crashed. National lost months worth of production and had to buy parts from competitors and remark them for sale.
It's also really interesting to read about the voltage limits. This explains why many low-power opamps don't go above 5.5-6V, even though it would be really useful if they could go just a bit above 4 AA batteries (6.4V).
One of the reasons chopper stabilized operational amplifiers are so common on low voltage CMOS *digital* processes is that there is no alternative on these processes for precision parts. Even Microchip Technologies makes them. And being made on a digital process makes them very inexpensive. But in the case of chopper stabilized amplifiers, you also get their disadvantages like high broadband noise, charge injection, and horrible overload recovery.
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Dave,
KOA Speer TLR2BDTD10L0F75
Res Metal Plate 1206 0.01 Ohm 1% 0.5W(1/2W) ±75ppm/C Pad SMD Automotive
±75ppm/C is not ±15ppm/C but 0.28 vs $4 for Vishay Y14870R01000B9R
Digikey $0.284/1@1000 us$ stock
Mouser $0.284/1@1000 stock
Searched using octopart dot com
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±75ppm/C is not ±15ppm/C but 0.28 vs $4 for Vishay Y14870R01000B9R
Vishay is 0.1% part. I think the whole point of using precise parts was to avoid calibration.
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The art of reading and understanding data sheets is a skill which separates the better engineers from the just good.
But I understand that this is not a task everyone enjoys, and I've heard complaints many times that datasheet hunting is "tedious".
I also follow a similar approach to Dave's. But the very first box I tick before starting any search is the "In Stock".
It is frustrating to find your dream part only to realize that it is in the unobtanium status.
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Searching only for in stock parts helps to get more common parts. However one may also miss some parts as sometimes even common parts run out of stock. If available have the usually stocked or similar tag helps.
One problem with the search function at Digikey / Mouser is that they sometimes mix typical and maximum values.
The definition of what the typical TC is also seems to be different. Some manufacturers seem to be a little more optimistic than others. I would have expected something like the RMS value of the TCs found. Over time the process may change (usually improve) a little.
The idea with the µCurrent is to have precision parts and not adjustment, thus the high accuracy parts and OP with lowest possible offset.
For a high BW use the noise could become really important.
For high performance / lowest noise the resistor steps are quite large. At the low end of the ranges the noise gets increasingly problematic. However more resistors would increase the costs quite a bit with the high precession parts.
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Searching only for in stock parts helps to get more common parts. However one may also miss some parts as sometimes even common parts run out of stock. If available have the usually stocked or similar tag helps.
One problem with the search function at Digikey / Mouser is that they sometimes mix typical and maximum values.
The definition of what the typical TC is also seems to be different. Some manufacturers seem to be a little more optimistic than others. I would have expected something like the RMS value of the TCs found. Over time the process may change (usually improve) a little.
The idea with the µCurrent is to have precision parts and not adjustment, thus the high accuracy parts and OP with lowest possible offset.
For a high BW use the noise could become really important.
For high performance / lowest noise the resistor steps are quite large. At the low end of the ranges the noise gets increasingly problematic. However more resistors would increase the costs quite a bit with the high precession parts.
High dynamic range is also important for the µCurrent. Not sure if there is something clever that could be done here, for example a logarithmic mode? Is there such a thing as a log autozero amp...
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Is there such a thing as a log autozero amp...
I don't think so. I was looking hard for an accurate log amp, found none. They were also expensive.
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High dynamic range is also important for the µCurrent. Not sure if there is something clever that could be done here, for example a logarithmic mode? Is there such a thing as a log autozero amp...
Dynamic range on the order of 26 bits with reasonable accuracy is feasible with a logarithmic design. I do not remember ever seeing an automatic "zero" logarithmic converter but it could be done using a pair or more of reference currents.
Since the output would be a logarithm, I am not sure it makes sense without a dedicated meter to show units and I have seen current meters which did exact this to show picoamps to milliamps without range switching.
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High dynamic range is also important for the µCurrent. Not sure if there is something clever that could be done here, for example a logarithmic mode? Is there such a thing as a log autozero amp...
Dynamic range on the order of 26 bits with reasonable accuracy is feasible with a logarithmic design. I do not remember ever seeing an automatic "zero" logarithmic converter but it could be done using a pair or more of reference currents.
Since the output would be a logarithm, I am not sure it makes sense without a dedicated meter to show units and I have seen current meters which did exact this to show picoamps to milliamps without range switching.
A few months ago I built a design around TI’s LOG114 plus a programmable power supply. On soldered breadboard (Busboard with solid copper groundplane) I could reasonably resolve from 100pA to 5mA, so not at all far off 26 bits.
The problem with this approach is that it’s dependent on a virtual ground, and the device loses its accuracy beyond a few mA as the virtual ground is too weak.
I used an on board MCU and LCD display, and powered it with a battery. I used the MCU’s on chip 12 bit ADC.
You have to be very careful when measuring, I put the entire measuring device and DUT into a foil tray.
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High dynamic range is also important for the µCurrent. Not sure if there is something clever that could be done here, for example a logarithmic mode? Is there such a thing as a log autozero amp...
Dynamic range on the order of 26 bits with reasonable accuracy is feasible with a logarithmic design. I do not remember ever seeing an automatic "zero" logarithmic converter but it could be done using a pair or more of reference currents.
Since the output would be a logarithm, I am not sure it makes sense without a dedicated meter to show units and I have seen current meters which did exact this to show picoamps to milliamps without range switching.
The scale could be some number of millivolts per DbV, or something like that, so it would work with any multimeter or scope?
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High dynamic range is also important for the µCurrent. Not sure if there is something clever that could be done here, for example a logarithmic mode? Is there such a thing as a log autozero amp...
Dynamic range on the order of 26 bits with reasonable accuracy is feasible with a logarithmic design. I do not remember ever seeing an automatic "zero" logarithmic converter but it could be done using a pair or more of reference currents.
Since the output would be a logarithm, I am not sure it makes sense without a dedicated meter to show units and I have seen current meters which did exact this to show picoamps to milliamps without range switching.
A few months ago I built a design around TI’s LOG114 plus a programmable power supply. On soldered breadboard (Busboard with solid copper groundplane) I could reasonably resolve from 100pA to 5mA, so not at all far off 26 bits.
The problem with this approach is that it’s dependent on a virtual ground, and the device loses its accuracy beyond a few mA as the virtual ground is too weak.
I used an on board MCU and LCD display, and powered it with a battery. I used the MCU’s on chip 12 bit ADC.
You have to be very careful when measuring, I put the entire measuring device and DUT into a foil tray.
Looks like a good chip, could probably be pressed into service for a logarithmic uCurrent.
It would be awesome to lose the range switching problem altogether. Maybe it doesn't have to be as good as a real logarithmic response, perhaps a crude approximation with two selectable linear slopes is enough - the second slope goes into effect at some fixed output voltage, say 1V to keep things simple.
The second slope could then be x1, x10, x100, x1000 for outputs beyond +/- 1V
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Since the output would be a logarithm, I am not sure it makes sense without a dedicated meter to show units and I have seen current meters which did exact this to show picoamps to milliamps without range switching.
The scale could be some number of millivolts per DbV, or something like that, so it would work with any multimeter or scope?
I do not think that would be usable. The analog logarithmic current meters I have seen simply used a log scale on the meter and the same would apply to an oscilloscope; use a log scale for the vertical axis.
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Since the output would be a logarithm, I am not sure it makes sense without a dedicated meter to show units and I have seen current meters which did exact this to show picoamps to milliamps without range switching.
The scale could be some number of millivolts per DbV, or something like that, so it would work with any multimeter or scope?
I do not think that would be usable. The analog logarithmic current meters I have seen simply used a log scale on the meter and the same would apply to an oscilloscope; use a log scale for the vertical axis.
Example: we set 0 mA as the 0dB reference level at the middle of two log scales.
so
200dB = 100mA
180dB = 10mA
160dB = 1mA
140dB = 100uA
120dB = 10uA
100dB = 1uA
80dB = 100nA
60dB = 10nA
40dB = 1nA
20dB = 100pA
0dB = 10pA or less (uCurrent can't go that low anyway)
-20dB = -100pA
-40dB = -1nA
-60dB = -10nA
-80dB = -100nA
-100dB = -1uA
-120dB = -10uA
-140dB = -100uA
-160dB = -1mA
-180dB = -10mA
-200dB = -100mA
So the output of the uCurrent could vary from +2000mV to -2000mV to represent the current magnitude and direction of +200.0dB to -200.0dB (1 decimal point).
This could look good both on a scope and on a DMM.
10 decades of log amplifier might be a tall order... but the highest couple of ranges could be a "cheat" of some kind... and 0dB could be 100pA instead of 10pA, realistically. 8 decades doesn't seem too terrible?
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Analog logarithmic circuits usually rely on the diode IV curve and are quite a bit temperature dependent. If the measured current directly goes through a diode, there would be self heating at the higher ranges (e.g. > 100 µA).
Even with some compensation it would not be very accurate. One has some 60 mV for 20 dB and 2 mV/K of temperature drift. So even of 90 % are compensated it would be some 0.05 dB/K of residual temperature effect.
Drift of the OP would be the smallest problem. Temperature regulation needs to much power to practically used battery power.
Due to the large drop with a diode it would need some kind of logarithmic trans-impedance amplifier, so only for currents the amplifier can drive, so maybe up to some 10 mA.
It would be a completely different instrument / tool, and not really workung well with negative currents, or just zero.
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I wonder if an oven could solve the problem. Like, a log-amp in LM399 package with an integrated heater.
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A log amp with an oven is a real option. It would at least avoid the temperature drift compensation and variable scaling factor. I have seen such plans using a quad transistor array: 2 heaters, one sensor and one as a diode for the log function. One may be able to use an LM723 (the main amplifier as a heater, the output transistor or auxiliary zener as sensor and the transistor for current limiting as the log element.
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A log amp with an oven is a real option. It would at least avoid the temperature drift compensation and variable scaling factor. I have seen such plans using a quad transistor array: 2 heaters, one sensor and one as a diode for the log function. One may be able to use an LM723 (the main amplifier as a heater, the output transistor or auxiliary zener as sensor and the transistor for current limiting as the log element.
I had no idea that log amplifiers were so sensitive to temperature, but it kind of does make sense. We are dealing with something like 24 bits of solution to span 8 decades, which is probably always going to be "challenging" to get excellent results from.
Perhaps there is a simple way to approximate a log response with a few linear segments, which would be "good enough for Australia" for this application?
(Love the LM723 idea!)
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Another option for a log amplifier might be to use a variable gain amplifier.
For example, this snippet from the Texas Instruments VCA810 data sheet:
[attachimg=1]
Maybe the OPA820 op amp could be replaced with an auto-zero op amp for DC applications?
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The problem with the log amplifier is not so much the large dynamic range, but the inherent temperature dependence. The scale factor is kT/e and thus proportional to abs temperature. Something like an PT1000 can give an approximate compensation, but not perfect.
There are reasonable cheap resistor arrays (e.g. MMPQ3904) available and the LM723 is TO99 is rare - AFAIK it also lacks the auxiliary zener. For the simple log circuit there is no need to have matching.
The trick would be using the nonlinear diode / transistor directly in the input stage as a shunt replacement. If one still has a shunt resistor to start with, one has the limited dynamic range from the small drop at the shunt.
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The problem with the log amplifier is not so much the large dynamic range, but the inherent temperature dependence. The scale factor is kT/e and thus proportional to abs temperature. Something like an PT1000 can give an approximate compensation, but not perfect.
There are reasonable cheap resistor arrays (e.g. MMPQ3904) available and the LM723 is TO99 is rare - AFAIK it also lacks the auxiliary zener. For the simple log circuit there is no need to have matching.
The trick would be using the nonlinear diode / transistor directly in the input stage as a shunt replacement. If one still has a shunt resistor to start with, one has the limited dynamic range from the small drop at the shunt.
The downside with that approach is that it doesn't start to work until the burden voltage is > 0.6V or something like that... and the whole point is to get the burden voltage down very low. Perhaps diodes / transistors can be placed somewhere else, e.g. in the feedback path?
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Because of the high drop the diode would have to be in a feedback path, a little like a trans-impedance amplifier.
This a quite common configuration for a log amplifier. The downside is that the current range is a little limited to maybe some 10 mA or so, because of heating in the transistor, even with a low voltage (e.g. some 0.5 V).
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The emitter resistance of the transistor limits accuracy at high currents before self heating by adding to the Vbe. The collector voltage is very low limiting power dissipation. At low currents leakage limits accuracy but picofarad measurements are possible.
Note that the transistor connection yields several more decades of dynamic range compared to the diode connection; ignore the limitations of diode connected logarithmic amplifiers.
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The transistor(s) could perhaps be switched in and out of the feedback by means of another switch on the uCurrent - so the device can retain its three "standard" linear ranges, and add a Logarithmic mode for those situations that call for it.
Winner, winner, chicken dinner! :D
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The logarithmic part would be essentially independent. In addition the log part would need quite some current, especially when using temperature stabilization. Even without, the measured current would have to come from the source.
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Dedicated log amps like the LOG114 and AD's ADL5303/5304 use trimmed internal thermal compensation.
Despite having the Eval boards for all of these devices, it was a bit of an uphill struggle to get them to work in the way I wanted, because they're really designed for photo diode applications. The TI LOG114 offered the path of least resistance in the end. It was surprisingly easy to get 100pA resolution in my own design. Because of the logarithmic nature of the device, if you're OK with nearly 3 sig figs of resolution across the range, you can make do with an MCU's 12 bit ADC. From grim experience, this is much easier than trying to get decent readings out of a 24 bit ADC, which will be much slower, and require averaging and oversampling.
You have to remember that you need to have a means of combining the log amp with a power supply of some sort. In its simplest form, you can use the on chip reference, but I used the MCU's DAC output with a bipolar voltage follower. In retrospect, bearing in mind it's only a few mA, you could use an op amp as a buffer, but keep in mind the limitations of any capacitive loading.
If you have a scope with a math exp function you can usefully probe the log amp's output (DS1000Z even has it), pretty darned useful for my use case which is button cell powered MCU based devices.
The biggest problem as a practical device for my use case is the weak floating ground which becomes a problem when you start going over a couple of mA, a real problem if you have inrush current or activity spikes because effectively your DUT's power supply voltage lags badly.
I've gone back to using 24 bits ADCs and some of the newer low offset current sense amps for current measurement for now, but practically speaking I'm finding it difficult to achieve much beyond 5.5 decades with an acceptable update rate, but it's a work in progress.
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A further complication is that the uCurrent has to be able to deal with both positive and negative currents, so it isn't strictly speaking enough with a log transfer function...
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A further complication is that the uCurrent has to be able to deal with both positive and negative currents, so it isn't strictly speaking enough with a log transfer function...
That’s right, although I’ve never meant to use the uCurrent in that mode.
Its only flaw in my use cases is that it lacks dynamic range within the limitations of the cobbled together systems I’ve come up with that use it, partially due to the instruments, and partially due to the uCurrent’s shunt resistor plus unipolar output.
To measure the kind of high dynamic ranges typical of today’s low duty cycle battery powered devices, 6+ decades of current measurement is very useful, with a top end ITRO 100mA and bottom end resolution in 100s of nA.
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If it is acceptable to deviate a little from the ideal bipolar logarithmic response, it seems possible to get very good results with the kind of relatively crude circuit below. The circuit gets pretty close to the "mathematically ideal" log response despite being an approximation "mix" of two lines and a log.
Here, the op amp handles the response near zero so performance is stable with no temperature effects there. The transistor temperature effects only come into play at high readings - and since they are linearized (a little!) with the emitter resistor, the effects are not really terrible at all, within a range of about 10C around room temperature.
Since the op amp solely determines the precision near zero, the circuit should work over many decades, perhaps as many as 8 or even more, depending on noise etc.?
[attachimg=1]
[attachimg=2]
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Since the op amp solely determines the precision near zero, the circuit should work over many decades, perhaps as many as 8 or even more, depending on noise etc.?
Without temperature compensation I expect error to be quite big. Although, may be a periodic auto calibration can be an option.
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Since the op amp solely determines the precision near zero, the circuit should work over many decades, perhaps as many as 8 or even more, depending on noise etc.?
Without temperature compensation I expect error to be quite big. Although, may be a periodic auto calibration can be an option.
In the TI (Burr Brown) LOG114 I've used, it has a parallel path, one for the DUT and the other with a precision current reference (implemented with an internal voltage ref plus external precision resistor), and then an internal difference amplifier. The two paths use internal matched transistors and thermally dependent resistors on each path to cancel out temperature drift.
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Since the op amp solely determines the precision near zero, the circuit should work over many decades, perhaps as many as 8 or even more, depending on noise etc.?
Without temperature compensation I expect error to be quite big. Although, may be a periodic auto calibration can be an option.
Here is what happens at temperatures of 15C and 25C (both green curves). The temperature only has an effect at higher readings (where the transistors are fully active) and looks like about a 2.5% error. The temperature error doesn't look much worse than the error made by approximating the log function in the first place...
Close enough for Australia? ...you can always switch back to a Linear mode on the uCurrent for precision measurements. The log function is really only to be able to "see" the wide dynamic signal on the scope, not necessarily to make precision measurements on it (how would you even do that). No?
[attachimg=1]
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The log function is really only to be able to "see" the wide dynamic signal on the scope, not necessarily to make precision measurements on it (how would you even do that).
I think it's a nice idea to have different tools with different trade-offs. Some people are happy with just measuring sleep current and active current separately, or even just measuring an average current. Others say they need to see transitions between the two or three states (active/sleep/wireless transmission) because this somehow affects battery efficiency due to battery chemistry or something (I'm clueless how this works).
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The log function is really only to be able to "see" the wide dynamic signal on the scope, not necessarily to make precision measurements on it (how would you even do that).
I think it's a nice idea to have different tools with different trade-offs. Some people are happy with just measuring sleep current and active current separately, or even just measuring an average current. Others say they need to see transitions between the two or three states (active/sleep/wireless transmission) because this somehow affects battery efficiency due to battery chemistry or something (I'm clueless how this works).
You just pretty much exactly described my use case - keeping an eye on transitions between active/sleep/wireless transmission. I found the µCurrent simply doesn't have enough dynamic range to do that without running multiple tests at different ranges. Of course, you could always buy a second µCurrent, put both in series, set them to different ranges, and feed their outputs into separate channels on your scope! :D
As other posters have pointed out, a high quality log function is unlikely to be simple or cheap enough for a relatively inexpensive device like the µCurrent. So we are looking at either auto-ranging (which some competitors to the µCurrent have implemented) or a very good but not perfect log function. Auto-ranging is much more likely to introduce weird artifacts in the results than a log function, as the device hops between ranges "chasing" the signal...
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A further complication is that the uCurrent has to be able to deal with both positive and negative currents, so it isn't strictly speaking enough with a log transfer function...
I think there is a Burr-Brown or Analog Devices application note which solved this problem with a translinear current inverter which has more dynamic range than an inverting amplifier.
I remember how this works now and it applies especially to integrated log amplifiers.
The problem was how to accept a positive current into the common NPN log amplifier which only works with negative currents. An inverting current amplifier using a transimpedance amplifier followed by a resistor will not work because it does not have enough dynamic range. A current mirror would destroy the virtual ground.
The solution is to add an operational amplifier to a current mirror to make the input into a virtual ground, which results in a log and antilog pair. It is the same circuit as the transimpedance amplifier followed by a resistor except the two resistors are replaced with diodes or transistors. The diodes or transistors compensate each other for temperature so no special requirements other than matched devices are required.
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A further complication is that the uCurrent has to be able to deal with both positive and negative currents, so it isn't strictly speaking enough with a log transfer function...
I think there is a Burr-Brown or Analog Devices application note which solved this problem with a translinear current inverter which has more dynamic range than an inverting amplifier.
I remember how this works now and it applies especially to integrated log amplifiers.
The problem was how to accept a positive current into the common NPN log amplifier which only works with negative currents. An inverting current amplifier using a transimpedance amplifier followed by a resistor will not work because it does not have enough dynamic range. A current mirror would destroy the virtual ground.
The solution is to add an operational amplifier to a current mirror to make the input into a virtual ground, which results in a log and antilog pair. It is the same circuit as the transimpedance amplifier followed by a resistor except the two resistors are replaced with diodes or transistors. The diodes or transistors compensate each other for temperature so no special requirements other than matched devices are required.
Would it be difficult to implement this 'on the cheap' outside the confines of an integrated circuit?
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Would it be difficult to implement this 'on the cheap' outside the confines of an integrated circuit?
In the original context, it was always implemented with discrete parts so that an NPN log amplifier on an integrated circuit could accept positive inputs. It helps to make up for the lack of good PNP matched pairs.
But the requirements are the same as any log amplifier except that temperature compensation is not required; it requires a precision operational amplifier and a matched diodes or transistor. I assume that a translinear version using the transistors as transistors instead of diodes would perform better but maybe not since in this case, the extra errors from diode operation cancel out.
Now whether that is the best way to implement a bipolar input log amplifier, I do not know. Maybe it is better to just implement two complete log amplifiers with opposite polarity although this requires a good PNP matched pair for the positive input.
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I found another example where the circuit idea I mentioned was used. Check out figure 9 on PDF page 5 of PMI (Precision Monolithic Incorporated) application note 106 which is now available as an Analog Devices application note with PMI references scrubbed:
https://www.analog.com/media/en/technical-documentation/application-notes/28080533AN106.pdf (https://www.analog.com/media/en/technical-documentation/application-notes/28080533AN106.pdf)
The circuit itself may be useful to some people for current measurement but note that it is *not* logarithm responding. On the other hand, no temperature compensation is needed as with a square and square root circuits based on the same idea.
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I found another example where the circuit idea I mentioned was used. Check out figure 9 on PDF page 5 of PMI (Precision Monolithic Incorporated) application note 106 which is now available as an Analog Devices application note with PMI references scrubbed:
https://www.analog.com/media/en/technical-documentation/application-notes/28080533AN106.pdf (https://www.analog.com/media/en/technical-documentation/application-notes/28080533AN106.pdf)
The circuit itself may be useful to some people for current measurement but note that it is *not* logarithm responding. On the other hand, no temperature compensation is needed as with a square and square root circuits based on the same idea.
Nice app note - I love that era of analog circuitry...
[edit] I have to think a little about the Figure 9 circuit, it is kind of cool...
The circuit in Figure 25 is more like what I was trying to do earlier, and I was also trying to partially use the logarithmic capabilities of the P/N junctions...
I think an approximation like that with 2 or 3 lines could work very well, and also be easy to understand on a linear (non-logarithmic) scope display... like instant automatic range switching.
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I have to think a little about the Figure 9 circuit, it is kind of cool...
I always have to study it again when I encounter it but it gets easier each time. I do not work with translinear circuits often enough to be comfortable with them.
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Join the chorus in praising this old app note.
And one knows it is old when it shows both a NAB and RIAA amplifiers!
I ignore whether this level of analog engineering proficiency is still actively growing, or will it die when the old beards start retiring. I suspect the latter.
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Join the chorus in praising this old app note.
And one knows it is old when it shows both a NAB and RIAA amplifiers!
I ignore whether this level of analog engineering proficiency is still actively growing, or will it die when the old beards start retiring. I suspect the latter.
The essential thoughts (or "transfer functions" of the analog circuits, if you like) are still with us, whether implemented as software algorithms (e.g. in a DSP) or implemented by taking advantage of the behaviour of fundamental components.
What I admire about the old style analog designs is how much complexity can be covered by the natural behaviour of a handful of basic components...
Even today: if you want to make something with a microprocessor, it sometimes helps to process the signal in the analog domain before converting to digital (e.g. to get away with a slower or less feature rich processor, for example)
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I found a temperature compensated log op amp circuit in "Operational Amplifiers 5:th Edition, G. Clayton, S. Winder, Newnes, 2003"
[attachimg=1]
Looks like the temperature compensating ability comes at a price: simulating it in LTSpice shows that it is only capable of about 3.5 decades of operation, which is significantly less than the unstabilized single transistor circuits.
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A further complication is that the uCurrent has to be able to deal with both positive and negative currents, so it isn't strictly speaking enough with a log transfer function...
Most people don't use the uCurrent in bipolar mode, so I'm thinking any new design might just drop that.
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±75ppm/C is not ±15ppm/C but 0.28 vs $4 for Vishay Y14870R01000B9R
Vishay is 0.1% part. I think the whole point of using precise parts was to avoid calibration.
Correct.
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A further complication is that the uCurrent has to be able to deal with both positive and negative currents, so it isn't strictly speaking enough with a log transfer function...
Most people don't use the uCurrent in bipolar mode, so I'm thinking any new design might just drop that.
Funny, the true AC capability is one of the things I like the most about it - I frequently use the uCurrent to measure small AC currents (and voltages, in low impedance circuits) where it works as an "oscilloscope booster" or "DMM booster" for frequencies below 250KHz. (Perhaps create a topic "Creative uses of the uCurrent", could be interesting and create some buzz?)
Even for the pulsating DC use case, sometimes the current might have unintentional excursions into the negative side due to the way the DUT (mis)behaves - which you would then not be able to see.
If the user reverses polarity on a unipolar device, there would now be no signal displayed - user can no longer tell the difference between 'no signal' and 'reverse polarity', which for sure would lead to some wasted time here and there.
To my mind, the bipolar capability of the original uCurrent was a stroke of genius that it would be sad to lose!
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To my mind, the bipolar capability of the original uCurrent was a stroke of genius that it would be sad to lose!
Noted!
Looking at using a 9V battery supply for the new design, so enough range to give decent bipolar supply.
Could do some sort of user switched split rail system to get the best of both worlds though.
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Even for the pulsating DC use case, sometimes the current might have unintentional excursions into the negative side due to the way the DUT (mis)behaves - which you would then not be able to see.
Allowing the virtual ground offset to be changed, as I've done, can give a larger positive range while still being able to see small negative swings.
Modification number 4:
https://www.eevblog.com/forum/crowd-funded-projects/current-gold-on-kickstarter/msg448208/#msg448208 (https://www.eevblog.com/forum/crowd-funded-projects/current-gold-on-kickstarter/msg448208/#msg448208)
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To my mind, the bipolar capability of the original uCurrent was a stroke of genius that it would be sad to lose!
Noted!
Looking at using a 9V battery supply for the new design, so enough range to give decent bipolar supply.
Could do some sort of user switched split rail system to get the best of both worlds though.
9V battery would be awesome, I go through a lot of CR2032 batteries - 50 hours sounds like a lot, but sometimes an experiment can run a long time and it becomes an issue. The extended range would be super useful too, of course. You probably wouldn't need a switchable split rail system with 9V because the overlap between the ranges would now be "big enough" to be able to find a home... But a switchable split rail would definitely be a cool thing for edge cases!
Another idea for the suggestion box: Quite frequently, I am trying to monitor small currents between large pulses. The uCurrent as it stands doesn't work so well for that, because the 10K shunt is in use to measure the small currents - sadly, the 10K resistance stops the DUT from being able to draw its occasional "breath" of high current, so it malfunctions.
A possible solution to that is to add a couple of (switchable) shunt by-pass diodes for the 10K shunt - perhaps a pair of low Vf Shottkys, or even transistors wired as diodes to minimize drop. That way, you can monitor ultra low currents over a long time period on your scope, and not have the DUT crash if it occasionally needs to inhale a large pulse of current.
I usually end up putting a diode externally to work around this problem. - Maybe something better than a diode could be used, to give a more clearly defined characteristic, but the ability to actually measure the high current pulse is not necessary. All we are looking to do is allow the DUT to keep working when drawing too much current for the nA range. After all, you can just switch to a higher range and measure the pulse there, if that's what you want to do.
[Edit: the 10 ohm shunt should probably also be included in this scheme, as it too is capable of dropping enough voltage during a pulse to worry the DUT]
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9V battery would be awesome, I go through a lot of CR2032 batteries - 50 hours sounds like a lot, but sometimes an experiment can run a long time and it becomes an issue. The extended range would be super useful too, of course. You probably wouldn't need a switchable split rail system with 9V because the overlap between the ranges would now be "big enough" to be able to find a home... But a switchable split rail would definitely be a cool thing for edge cases!
I was thinking about that earlier; instead of lower noise or higher precision, micropower operation might be more useful for extended operating time. Micropower chopper stabilized operational amplifiers are available but you have to specifically select them; just because something is CMOS does not make it low power. Bipolar precision parts may have an advantage here with lower noise and higher bandwidth.
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To my mind, the bipolar capability of the original uCurrent was a stroke of genius that it would be sad to lose!
Noted!
Looking at using a 9V battery supply for the new design, so enough range to give decent bipolar supply.
Could do some sort of user switched split rail system to get the best of both worlds though.
9V battery would be awesome, I go through a lot of CR2032 batteries - 50 hours sounds like a lot, but sometimes an experiment can run a long time and it becomes an issue. The extended range would be super useful too, of course. You probably wouldn't need a switchable split rail system with 9V because the overlap between the ranges would now be "big enough" to be able to find a home... But a switchable split rail would definitely be a cool thing for edge cases!
Another idea for the suggestion box: Quite frequently, I am trying to monitor small currents between large pulses. The uCurrent as it stands doesn't work so well for that, because the 10K shunt is in use to measure the small currents - sadly, the 10K resistance stops the DUT from being able to draw its occasional "breath" of high current, so it malfunctions.
A possible solution to that is to add a couple of (switchable) shunt by-pass diodes for the 10K shunt - perhaps a pair of low Vf Shottkys, or even transistors wired as diodes to minimize drop. That way, you can monitor ultra low currents over a long time period on your scope, and not have the DUT crash if it occasionally needs to inhale a large pulse of current.
I usually end up putting a diode externally to work around this problem. - Maybe something better than a diode could be used, to give a more clearly defined characteristic, but the ability to actually measure the high current pulse is not necessary. All we are looking to do is allow the DUT to keep working when drawing too much current for the nA range. After all, you can just switch to a higher range and measure the pulse there, if that's what you want to do.
My new design will have automatic range switching.
I don't know about having a long term battery just for the edge cases where you need it for long term data logging. Perhaps just an external DC jack for that? Wire in your own external battery if needed.
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My new design will have automatic range switching.
That sounds very cool! It would have to switch very fast to avoid interrupting the current. I imagine if the user selects a range manually, it could automatically "uprange" and immediately fall back to the user selected range again when the pulse is gone.
I'm struggling to see, though, how you would tell which range was in use, when you look at the data later? (i.e. the pulse would now look like it was part of the low current?) This might be hard to crack without a separate "range selected" terminal of some kind.
I don't know about having a long term battery just for the edge cases where you need it for long term data logging. Perhaps just an external DC jack for that? Wire in your own external battery if needed.
An external DC jack is a much better idea. ...Also, you can get rechargeable 9V Li-Ion batteries these days... perhaps the DC jack could optionally be a way to recharge the internal battery in the uCurrent? It is a little inconvenient to open and close with 4 screws every time you need to replace the battery when it is in heavy use...
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That sounds very cool! It would have to switch very fast to avoid interrupting the current. I imagine if the user selects a range manually, it could automatically "uprange" and immediately fall back to the user selected range again when the pulse is gone.
Yep, it'll do that.
I'm struggling to see, though, how you would tell which range was in use, when you look at the data later? (i.e. the pulse would now look like it was part of the low current?) This might be hard to crack without a separate "range selected" terminal of some kind.
I've got that covered ;D
An external DC jack is a much better idea. ...Also, you can get rechargeable 9V Li-Ion batteries these days... perhaps the DC jack could optionally be a way to recharge the internal battery in the uCurrent? It is a little inconvenient to open and close with 4 screws every time you need to replace the battery when it is in heavy use...
I'm weighing up the pro's and cons of a rechargeable solution. I like the simplicity of a 9V battery (or two for an ultra dynamic range mode). But yes, it needs to be changeable without screws into a plastic case.
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I'm weighing up the pro's and cons of a rechargeable solution. I like the simplicity of a 9V battery (or two for an ultra dynamic range mode). But yes, it needs to be changeable without screws into a plastic case.
I also like the 9v battery (or 2x 9v battery) idea. If the battery is accessible without opening the case, the uCurrent probably shouldn't get involved in battery charging at all. We can even live without a DC jack, as external power could be applied directly to the 9V battery connector(s) for long term data logging.
The user can then choose the battery chemistry that works for them, and whatever charging technology works with it.
Some rechargeable 9V batteries are a little larger than the standard ones... hopefully there is room for a "generous" battery compartment! :D
For rechargeable use, I was thinking of the li-ion versions of the venerable 9v battery - e.g. similar to these ones:
[attachimg=1]
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The problem with rechargeable versions of the standardized 9 volt battery is that there are so many different kinds.
Energy density is better with cylindrical cells if lower voltage is acceptable. I remember ads from either Maxim or Linear Technology about "how to suck a battery dry" which compared dual AA cells to 9 volts with various power management solutions.
Supply current for a pair of MAX4239s is 1.2 milliamps? That can be reduced to 1/10th pretty easily and an extreme design could be 1/50th. How much bandwidth is required when the measuring instrument is a multimeter?
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The problem with rechargeable versions of the standardized 9 volt battery is that there are so many different kinds.
Energy density is better with cylindrical cells if lower voltage is acceptable. I remember ads from either Maxim or Linear Technology about "how to suck a battery dry" which compared dual AA cells to 9 volts with various power management solutions.
Supply current for a pair of MAX4239s is 1.2 milliamps? That can be reduced to 1/10th pretty easily and an extreme design could be 1/50th. How much bandwidth is required when the measuring instrument is a multimeter?
The instrument is most often a 'scope, in my case at least. So the current uCurrent current [ :D ] bandwidth of about 250KHz is about right. Could live with a little less, but it is nice to see the "shape" of the current consumption.
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There are two uses:
1) measuring current with the scope
2) measuring current with a voltmeter with reduced burden voltage. With old DMMs this may add another current range. It may also improve on the choice of actual shunts in a newer meter - modern auto-scale meters may have a limited choice of shunts. The cases where the reduced burden is really needed are relatively rare. It may still be handy to simplify things.
For the DMM ind DC one would usually only need low BW, like some 100 Hz.
For the scope use one would like to have a relatively high BW, but also low noise. Especially with the scope the low offset is not really needed, low noise would normally be preferred. The same is true with AC measurements.
With the low burden voltage the dynamic range is naturally reduced. So the µC current is not the right starting point to measure the highly variable current of a µC. This would be another piece of equipment, e.g. using voltage regulation with a FET and measure the current at the other side of the FET.
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With the low burden voltage the dynamic range is naturally reduced. So the µC current is not the right starting point to measure the highly variable current of a µC. This would be another piece of equipment, e.g. using voltage regulation with a FET and measure the current at the other side of the FET.
I am struggling to understand. Isn't the current (and therefore also the dynamic range) the same on both sides of the FET, so the problem of measuring it is the same? (I do see that you could use a larger shunt resistor in front of the FET, but I don't see a change in dynamic range of the current)
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I'm weighing up the pro's and cons of a rechargeable solution. I like the simplicity of a 9V battery (or two for an ultra dynamic range mode). But yes, it needs to be changeable without screws into a plastic case.
I also like the 9v battery (or 2x 9v battery) idea. If the battery is accessible without opening the case, the uCurrent probably shouldn't get involved in battery charging at all.
I've got a neat idea for an external 9V battery.
We can even live without a DC jack, as external power could be applied directly to the 9V battery connector(s) for long term data logging.
For my idea the contacts would be external, so would still need a DC jack or screw terminals. I presume DC jack is preferable in that case?
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Supply current for a pair of MAX4239s is 1.2 milliamps? That can be reduced to 1/10th pretty easily and an extreme design could be 1/50th. How much bandwidth is required when the measuring instrument is a multimeter?
The instrument is most often a 'scope, in my case at least. So the current uCurrent current [ :D ] bandwidth of about 250KHz is about right. Could live with a little less, but it is nice to see the "shape" of the current consumption.
[/quote]
For the new design I'm targeting around 4-5 times that bandwidth, and lower noise. Comes at the expense of battery life though, hence the 9V vs coin cells.
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We can even live without a DC jack, as external power could be applied directly to the 9V battery connector(s) for long term data logging.
For my idea the contacts would be external, so would still need a DC jack or screw terminals. I presume DC jack is preferable in that case?
A DC jack is a nicer solution than screw terminals, especially if user has more than one external power source. For convenience, the unit could be supplied with a couple of spare DC plugs or even pigtails, so user can easily make custom cables for external power sources?
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Supply current for a pair of MAX4239s is 1.2 milliamps? That can be reduced to 1/10th pretty easily and an extreme design could be 1/50th. How much bandwidth is required when the measuring instrument is a multimeter?
The instrument is most often a 'scope, in my case at least. So the current uCurrent current [ :D ] bandwidth of about 250KHz is about right. Could live with a little less, but it is nice to see the "shape" of the current consumption.
For the new design I'm targeting around 4-5 times that bandwidth, and lower noise. Comes at the expense of battery life though, hence the 9V vs coin cells.
Larger bandwidth and lower noise is interesting but I'd rather "spend the improvement" on a 1000x amplification option, than increased bandwidth... There are times when 100x is not quite enough to overcome the low sensitivity of the scope input, if the current being measured is extremely small.
I have never actually encountered a situation where the bandwidth of the existing uCurrent was not large enough for its intended use. Most power rails have capacitors on them that would limit the bandwidth of the supply current to something lower than the uCurrent can already do, in any case?
That said, if there was a way to have cake and eat, I would soon think of a use for the extra bandwidth! :D Bode plotting, capacitor ESR testing, component testing, etc. etc.
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With the low burden voltage the dynamic range is naturally reduced. So the µC current is not the right starting point to measure the highly variable current of a µC. This would be another piece of equipment, e.g. using voltage regulation with a FET and measure the current at the other side of the FET.
I am struggling to understand. Isn't the current (and therefore also the dynamic range) the same on both sides of the FET, so the problem of measuring it is the same? (I do see that you could use a larger shunt resistor in front of the FET, but I don't see a change in dynamic range of the current)
The other side of the FET could use a larger shunt. It is the low voltage drop that limits the dynamic range.
For low currents a trans-impedance amplifier is a real option. One can get a low burden and large dynamic range. It also makes the protection easier.
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With the low burden voltage the dynamic range is naturally reduced. So the µC current is not the right starting point to measure the highly variable current of a µC. This would be another piece of equipment, e.g. using voltage regulation with a FET and measure the current at the other side of the FET.
I am struggling to understand. Isn't the current (and therefore also the dynamic range) the same on both sides of the FET, so the problem of measuring it is the same? (I do see that you could use a larger shunt resistor in front of the FET, but I don't see a change in dynamic range of the current)
The other side of the FET could use a larger shunt. It is the low voltage drop that limits the dynamic range.
For low currents a trans-impedance amplifier is a real option. One can get a low burden and large dynamic range. It also makes the protection easier.
Ah, the dynamic range of the voltage across the shunt is of course higher with the larger shunt resistor.... dooh, clearly I did not have enough coffee! :D
The concept of the uCurrent is to use an amplifier to make up for the fact that you are using a smaller shunt. As long as that doesn't become too noisy, we are in good shape? It makes the uCurrent more versatile that it is not part of a larger device or power supply.
I like the idea of a trans impedance amplifier, but now the properties of that amplifier becomes part of the circuit that it is connected to - so we may run into bandwidth issues etc? - with a shunt, the circuit still works even if the amplifier reading the shunt can't keep up, shifts phase, or whatever...
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The TIA would need a resistor at the input to isolate it from a capacitive input. The other point is to have a suitable parallel capacitor in the feedback to limit the BW to a little less than what the OP could provide. This should result in a reasonable well behaved impedance.
The TIA idea is good for the smaller currents (e.g. < 100 µA), so that the current from the OP does not have to be so large and one can tolerate some 100-1000 Ohms at the input for isolation. The shunt + amplifier solution is limited to small currents, as the shunt will have noise of it's own. So instead of a 10 K shunt, I would definitely prefer a TIA (e.g. with some 1 M in the feedback). The shunt + amplifier solution is limited by noise and the DC offset, especially with a small burden and using 1:1000 steps. With a maximum output of some +-2 V of the µCurrent one would have 20 mV max at the shunt and thus only some 20 µV at the shunt just before the step to the next larger shunt would be possible - this is really small with not much resolution left.
Even with a low drop, I am not so sure that automatic range switching is such a good idea. The change in shunt resistance will have an effect on the circuit and sometimes one would still need a manual mode to get the right range before an expected jump to a higher current.
To get more dynamic range for the output with a limited supply, one can use an active output a little like a bridge driven amplifier instead of a simple fixed virtual ground. So if the output is positive the other output terminal and input side virtual ground could be closer to the negative side. This could nearly double the range, though with a slight limit for fast transients. There is no need to switch manually, it can be done with a not so critical inverter circuit.
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For the new design I'm targeting around 4-5 times that bandwidth, and lower noise. Comes at the expense of battery life though, hence the 9V vs coin cells.
I wonder at what point a chopper stabilized low noise bipolar amplifier has a lower noise for the same power draw than only a chopper stabilized amplifier. Power draw was never a consideration when I designed them.
My intuition is that the flat low frequency noise of a chopper stabilized amplifier means that a micropower part could be used, and since bipolar parts have lower broadband noise for the same power, the combined noise would always be better. Plus only a single chopper stabilized part would be required to correct the first stage.
The only caveat is that for bandwidth comparable to CMOS parts which benefit from lower transconductance, decompensated bipolar operational amplifiers would be required which somewhat limits parts selection and even so is a trade off unless external compensation is supported which is practically unknown now.
So the ultimate low power low noise wide bandwidth design would use a discrete bipolar input stage, allowing for maximum decompensation, with low Rb transistors (1) in parallel with a micropower chopper stabilized amplifier. This would also have the benefit of allowing the input voltage noise and input current noise to be adjusted for lowest total noise given the source resistance of the input divider. Conveniently this also does away with the need for the most expensive high performance operational amplifiers.
(1) Expensive IC based transistors would be best here like the MAT series but cheap low noise audio transistors like the BC327/BC337 would also perform well. The various old Zetex "Super E Line" transistors currently made by Diodes, Inc. are likely even better but not characterized for this sort of application. Fairchild (bought by On Semiconductor) also makes these types of parts now.
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The TIA would need a resistor at the input to isolate it from a capacitive input. The other point is to have a suitable parallel capacitor in the feedback to limit the BW to a little less than what the OP could provide. This should result in a reasonable well behaved impedance.
The TIA idea is good for the smaller currents (e.g. < 100 µA), so that the current from the OP does not have to be so large and one can tolerate some 100-1000 Ohms at the input for isolation. The shunt + amplifier solution is limited to small currents, as the shunt will have noise of it's own. So instead of a 10 K shunt, I would definitely prefer a TIA (e.g. with some 1 M in the feedback). The shunt + amplifier solution is limited by noise and the DC offset, especially with a small burden and using 1:1000 steps. With a maximum output of some +-2 V of the µCurrent one would have 20 mV max at the shunt and thus only some 20 µV at the shunt just before the step to the next larger shunt would be possible - this is really small with not much resolution left.
Perhaps the entire configuration of the amplifier can switch between TIA and Shunt mode, for different ranges?
Dave is talking about using +/-4.5V or even +/-9V in the new design, which would help the shunt amplifier solution generally? Perhaps the DC power jack could accept 12VDC, now the situation is 1 decade better...
If the amplifier could be switched between x100 and x1000, the range steps could be made smaller as well...
So many engineering compromises/design decisions to think about! :D
Even with a low drop, I am not so sure that automatic range switching is such a good idea. The change in shunt resistance will have an effect on the circuit and sometimes one would still need a manual mode to get the right range before an expected jump to a higher current.
Ideally there would not be any range switching at all, but that would require a logarithmic response... which seems surprisingly difficult, there isn't a simple/clean solution (other than an oven, which might be an energy pig) and even if the problems are resolved, it is harder to interpret the numbers afterwards...
Really we can live with selecting a range in advance, but somehow we have to dynamically "short circuit" the shunt if the DUT overdraws current for that range. I have used the "old" uCurrent with a diode across it to handle shunt overload, that worked well enough that it seems credible that range switching can be made to work - as long as it works as fast as a diode across the shunt would.
To get more dynamic range for the output with a limited supply, one can use an active output a little like a bridge driven amplifier instead of a simple fixed virtual ground. So if the output is positive the other output terminal and input side virtual ground could be closer to the negative side. This could nearly double the range, though with a slight limit for fast transients. There is no need to switch manually, it can be done with a not so critical inverter circuit.
That's a great idea - One problem is that the negative output would then float with respect to ground - sometimes, both the DUT and the measuring device (scope) are connected to the same ground. It would work great with a floating instrument or a floating source, of course. Perhaps it could be made a switchable x2 range extension feature, where the negative is plain ground unless range extension is engaged. With the 9V battery it would be an amazing range! :D