An Experimental AC Voltage Calibrator

[enut11]

Experimental AC Voltage Calibrator
I debated whether this project fits in the ‘Projects, Designs and Technical Stuff’ thread or the ‘Metrology’ one. Since I love precision electronics, here goes.

After spending a brief session with a Fluke 5200A AC Voltage Calibrator, I realised what a handy instrument it is. The 5200A offers a wide range of accurate high resolution voltage at frequency settings. Biggest drawbacks for me: 1) Bulk: The 5200 is a huge and heavy box that needs lots of bench space, 2) Cost: around $2000 USD or more for a good one.

So, whenever I come across an idea for a DIY AC Voltage Calibrator, it gets my attention. Recently, a friend suggested an interim solution. Not a ‘proper’ calibrator but a stable, variable amplitude/frequency source of AC volts. Such an  instrument, together with an accurate DMM, could be used to ‘transfer’ an AC voltage/frequency combo to another instrument.

The concept
Take a stable low distortion audio frequency generator and feed the signal into a HiFi audio amplifier. This alone will give you up to 20vRMS in the range of 20Hz-20KHz and possibly well beyond. For higher output voltages, connecting a 100v/8ohm audio line transformer, back-to-front, to the amplifier will give you up to 200vAC.

A simple concept, but does it work? How stable would the output be? These are the questions I needed to answer for myself. More to come…
enut11

[Jester]

I made something  like this quite a while ago before I had an ac calibrator. I used HP sig generator as source driving audio amp and conventional transformer running at about half its voltage rating. It’s a lot better than using MAINS voltage and capturing and correlating two sets of measurements, at least the measurements are relatively stable.

The bulky size, cost and cost of the conventional ac calibrators was also a deterrent for me. I prefer to have all the instruments that I use regularly within reach at my bench. Bottom row 2nd from right.

Clarke Hess (smaller less expensive ac calibrator) show up on eBay once in a while, I picked one up in non working order for next to nothing and fixed it, pretty decent little calibrator if super accuracy is not required. I tried to get a schematic from Clarke Hess, they were uncooperative. Troubleshooting a bad one without the schematic is a bit tricky. Mine just had a leaky diode.

[mzzj]

I did take a part in DMM intercomparison (8½ digit Fluke 8508A) with shoestring budget equipment some years ago.
Laptop soundcard, audio amplifier and bunch of transformers to get up to 700Vac 
Results looked good with approx 0,03% uncertainty up to amplifier max voltage.
transformer coupling increased the uncertainty to 0.06% at 700 Vac (3458A as reference is not that great here either)

Laptop sound card was much better choice than either of tested (digital) function generators.

[mawyatt]

Interesting idea. How stable is the audio amp gain and how flat over frequency?

A very simple and useful low voltage AC calibrator we used some time ago is based on a CD4060 divider chip and a 4.096MHz crystal. The VDD voltage for the CD4060 is created with a stable 5 volt reference, since the CD4060 is CMOS, the selected frequency divided output will swing from ground to ~VDD without any load (typical DMM isn't much of a load). The output voltage is an accurate 5.000 Vpp 50% duty cycle squarewave with an AC rms and DC average value of VDD/2 or 2.500 Volts and stable.

Of course the squarewave has harmonics and the frequency range must be taken into consideration, but works quite well on the various DMMs we have, KS34465A, HP34401A, AG34401A, SDM3065X, and DMM6500 just to verify low voltage AC performance.

Edit: Just plugged in the reference and let it run for a few minutes, it's AC reading on DMM6500 is 2.499264V RMS with 30uv SD. Also reads 2.500049V DC with 10uv SD.

Edit: Found the calibrator thread.
https://www.eevblog.com/forum/projects/diy-dvm-simple-dvm-cal-device/msg3157650/#msg3157650

Best,

[bdunham7]

The concept
Take a stable low distortion audio frequency generator and feed the signal into a HiFi audio amplifier. This alone will give you up to 20vRMS in the range of 20Hz-20KHz and possibly well beyond. For higher output voltages, connecting a 100v/8ohm audio line transformer, back-to-front, to the amplifier will give you up to 200vAC.

A simple concept, but does it work? How stable would the output be? These are the questions I needed to answer for myself. More to come…

Having done both (but not the transformer part) I can tell you that it works fine as long as you have a known good audio amplifier, an ordinary AWG and a good reference meter.  The accuracy limits of this setup will likely be determined by the reference meter.  A transformer will likely work as long as you stay well within its saturation limits.  I use one or two (bridged) Marantz monoblock amplifiers and I can do better than 80V_rms @ 50kHz.

The 5200A has significant limitations and having had/fixed/used them I don't miss having one.  The frequency controls have fine resolution, but poor accuracy--you need to use an external AWG to control them if you want precise frequency.  They only go to 120V_rms on their own, and that only up to 80kHz or so.  The frequency range goes to 1.2MHz, but that is only at a low voltage.  You can get those low-cost add-on amplifiers for your AWG that will do nearly as well if you need to go beyond what your audio amp can do.

[mawyatt]

You can get those low-cost add-on amplifiers for your AWG that will do nearly as well if you need to go beyond what your audio amp can do.

We designed and couple as AWG Buffer amps as supplements to the Juntek DPA-2698 and DPA-1698, one we built so far is based upon the HV OPA462 Op-Amp discussed here:

https://www.eevblog.com/forum/testgear/hv-buffer-amp-for-awg/msg3623953/#msg3623953

As a quick test, just plugged in the 2.5V RMS AC calibrator mentioned above and connected it to the OPA462 based AWG ~10X Buffer Amp.

The DMM6500 reads ~25.00667V, KS34465A 25.0085V, HP34401A 25.0558V, and AG34401A 25.0419V.
Then connected up the AWG at:
50Hz with 50.0550, 50.0554, 50.0245, and 49.9998 respectively
100Hz with 50.0460, 50.0437, 50.0215 and 49.9920
1KHz   with 50.0475, 50.0480, 50.0345, and 50.0030
10KHz with  50.0775, 50.0740, 50.0620, and 50.0330
50KHz with 49.9135, 49.9048, 49.8860, 49.8350

Haven't tried driving a transformer tho for higher voltage, that would be interesting to see how well these relative readings hold up across the various DMMs at higher voltages.

Edit: here's the AWG 10X Amp.

Best,

[enut11]

Well, some interesting ideas and suggestions already. That is why this is a great Forum for electronics enthusiasts. For now I will continue with my story and address any issues later.

I had a surplus custom instrument on hand with all the parts inside for a mono audio amplifier. It was based on the LM1875 20W audio amp chip and, with the included power transformer (36v CT), was able to give around 15vRMS output when fed with a 1KHz sinewave at the input. This was a good start. Using this setup I could generate AC voltages from 10Hz to 50KHz.

The next step was to find a suitable audio transformer for higher output voltages. Well, this was a learning experience in itself. High quality audio output transformers can cost many thousands of dollars! No doubt well worth it to the golden-ears brigade. For my purpose, I wanted to keep the cost down to well below a 5200A. So, for the initial proof-of-concept, I settled on a Jaycar Electronics (Australia)  MM1900 for less than $11.

I might add that audio output transformers are better suited for this role because of the extended frequency response due to the special steel laminations.

[enut11]

The signal generator
The output of open loop systems is very dependent on the quality of individual interconnected components and this project was no exception.

I started out by analysing the amplitude stability of my signal generators.
1) Leader LAG126S audio frequency
2) Feeltech model FY2300
3) Wavetek Model 80
4) Chinese kit - 'Low Distortion Audio Range Oscillator'

The best turned out to be the last one. The design is based on a simple state variable design with FET output control. Using a regulated +/-15v power supply this oscillator produces a clean low noise signal. Even so, the plot below shows there is a 60min warmup period before it stabilises to the point that it can be relied on for making good measurements.

Stated distortion is 0.006%.
I built a 1KHz + 10KHz kit and may even look at a variable frequency version.

[mawyatt]

Is it necessary to have a low distortion sine-wave for checking AC meter performance? As mentioned we used a square-wave since producing and accurate squarewave is straightforward using CMOS logic and flip-flops.

Seems that as long as the distortion products don't extend beyond the measuring instrument meter bandwidth, that they get accurately accounted for in the RMS computations, or in the analog RMS converter chip, which ever the meter utilizes.

Might expect some differences with meters that use different RMS methods, like computational or analog, but not so much between "like" technique instruments, again as long as the BW isn't pushed.

Curious as to what others think about this?

Best,

[enut11]

The amplifier
The next item to check was the amp itself. All AB class amplifiers dissipate several watts, even doing nothing, in order to produce a low distortion output. So, even though I was only going to amplify volts at very low currents, the standing amp bias current necessitated a large heat-sink and even a cooling fan.

National specify the following for LM1875 amp:
The LM1875 is a monolithic power amplifier offering up to 30 Watts Output Power very low distortion and high quality performance for AVO Typically 90 dB consumer audio applications. Low Distortion: 0.015%, 1 kHz, 20 W The LM1875 delivers 20 watts into a 4Ω or 8Ω load. Wide Power Bandwidth: 70 kHz on ±25V supplies. Using an 8Ω load and ±30V
Protection for AC and DC Short Circuits to supplies, over 30 watts of power may be delivered.

My noise measurements with a shorted input show about 30uV PP.

[enut11]

Is it necessary to have a low distortion sine-wave for checking AC meter performance? As mentioned we used a square-wave since producing and accurate squarewave is straightforward using CMOS logic and flip-flops.

Seems that as long as the distortion products don't extend beyond the measuring instrument meter bandwidth, that they get accurately accounted for in the RMS computations, or in the analog RMS converter chip, which ever the meter utilizes.

Might expect some differences with meters that use different RMS methods, like computational or analog, but not so much between "like" technique instruments, again as long as the BW isn't pushed.

Curious as to what others think about this?

Best,

My view is that square waves are only good for low frequency tests (<1KHz). Clean sine waves are needed for the higher frequencies .
enut11

[bdunham7]

Is it necessary to have a low distortion sine-wave for checking AC meter performance? As mentioned we used a square-wave since producing and accurate squarewave is straightforward using CMOS logic and flip-flops.

Seems that as long as the distortion products don't extend beyond the measuring instrument meter bandwidth, that they get accurately accounted for in the RMS computations, or in the analog RMS converter chip, which ever the meter utilizes.

For a basic check perhaps not, but for calibration to within the meters specs, you probably need a low-THD sine as specified.

One thing to think about is that in addition to any error that might occur simply because of dissimilar RMS conversion of non-sinusoids, the meters have a frequency-dependent accuracy spec.  Higher frequencies result in wider tolerances.  Take your square wave calibrator and look at how much energy there is in those upper harmonics and which tolerance bin they fall into.  Say you have a meter with a specified accuracy of +/- 0.02% @ 1kHz, but +/- 3% @ 50kHz.  What percentage of the signal is in that 49th or 51st harmonic?  How about the percentage of the signal that is in all the harmonics above 50kHz?  And how does the 3% error multiplied by that  percentage compare with the 0.02% error at the fundamental? 

Now if you are talking about two sine stimuli, one with 1% THD and the other with 0.1% THD, both primarily in the 2nd and 3rd harmonics, then perhaps the difference will be less.  This would not be difficult to test.  Perhaps the next time I'm working on a meter or a calibrator, I'll try it.

[alm]

Is it necessary to have a low distortion sine-wave for checking AC meter performance? As mentioned we used a square-wave since producing and accurate squarewave is straightforward using CMOS logic and flip-flops.

Seems that as long as the distortion products don't extend beyond the measuring instrument meter bandwidth, that they get accurately accounted for in the RMS computations, or in the analog RMS converter chip, which ever the meter utilizes.

Might expect some differences with meters that use different RMS methods, like computational or analog, but not so much between "like" technique instruments, again as long as the BW isn't pushed.

I don't think DMMs are generally specified for square(ish) wave signals. I would definitely want to use a bandwidth-limited signal limited well below the bandwidth of all meters involved. Then I imagine would kind of work, but I'd expect the uncertainty of your comparison to be higher than with sine waves due to the different frequencies involved and the finite attenuation of any filter.

Performance verification generally calls for frequencies across the bandwidth. For example for the HPAK 34401A, they use frequencies ranging from 20 Hz to 300 kHz (there's a reason why the Fluke 5200A has such a wide bandwidth). How would you verify the high end. A 300 kHz square wave? A 100 kHz square wave limited to 300 kHz (won't be very square)?

[enut11]

Line Output Transformer
The next step was to introduce the line output transformer to generate voltages  beyond the capability of the basic audio amplifier - up to 200vAC RMS.
This was my first encounter with audio transformers so was unsure what to expect. The transformer used for these tests is shown below. It is a Jaycar Electronics MM1900 100v/8ohm line speaker transformer with 'primary' tappings for 0.5W to 5W. The latter proved useful in setting the transformer ‘gain’.

The transformer 8ohm 'secondary' was connected to the ‘speaker’ output of the amp and the high impedance ‘primary’ connected to a DMM. For the MM1900, the gain was found to be x35 on the 2W tap so 200vAC out needed less than 6vAC from the audio amp which in this case had a 20:1 gain. This means that the signal generator only needed to supply less than 300mV to the amp.

I decided to test at a reduced output of 100vAC and the results appear below. Again, for about 60min, the output drops steadily, mainly due to the signal generator. After that, the 100vAC 1KHz signal proved to be surprisingly stable with relatively low noise. The second graph below shows the estimated noise in the flat part of the curve of approx 1400uV PP which equates to 14 PPM for a 100v signal. Not too bad for such a simple setup.

Frequency response of the $11 audio transformer was somewhat less than the amp, 50Hz-15KHz, but still useful.

[enut11]

As added insurance, I decided to regulate the LM1875 amp power supplies using LM317/337 chips. The standing amp bias was around 50mA. The unregulated supplies were +/-25v so I set the regulators to +/-20v. This had the effect of reducing the amp output to about 13v before clipping. Still plenty enough to drive the audio transformer.

What's next? More output of course!
My friend suggested a better audio amp and a coupled pair of output transformers. Australia’s Silicon Chip Mag (Dec 2021) have come up with a brilliant ‘Hummingbird’ design based on a very compact PCB (64mmx88mm). Specs include <0.008% distortion and 60 watts RMS into 8 ohms (22Vrms). Noise is down at the -118dB level. The amp has a gain of x18 and a frequency response of 1Hz to 150KHz!
The existing low distortion audio generator (<0.003% THD) should be ok but would probably benefit from being in a bigger, properly shielded metal box.

The output transformer: Altronics (Australia) have a line of quality audio output transformers with a  better frequency response. I have ordered two units (catalog M1120) with a view to coupling them in such a way as to share the output voltage – up to 300vAC should be possible without straining the transformer primaries. The M1120 are 20 watt units with a stated response of 30Hz-20KHz. They also have 4, 8 and 16ohm output taps which will come in handy when selecting the optimum ‘gain’.
enut11

[bdunham7]

As far as audio transformers go, units made for single-ended tube outputs may work well because they are air-gapped and designed not to saturate.  That and they may be just fine with 300V+ outputs.  The PA ones are a little lower caliber.

[David Hess]

Interesting idea. How stable is the audio amp gain and how flat over frequency?

It might be stable, but a typical amplifier will not be flat even to 0.1% over a significant frequency range.  Better accuracy will require leveling.

A very simple and useful low voltage AC calibrator we used some time ago is based on a CD4060 divider chip and a 4.096MHz crystal. The VDD voltage for the CD4060 is created with a stable 5 volt reference, since the CD4060 is CMOS, the selected frequency divided output will swing from ground to ~VDD without any load (typical DMM isn't much of a load). The output voltage is an accurate 5.000 Vpp 50% duty cycle squarewave with an AC rms and DC average value of VDD/2 or 2.500 Volts and stable.

That is how I have done it in the past to at least 0.02%.  I am not sure how much better because I do not have anything I trust that is better than that.

Of course the squarewave has harmonics and the frequency range must be taken into consideration, but works quite well on the various DMMs we have, KS34465A, HP34401A, AG34401A, SDM3065X, and DMM6500 just to verify low voltage AC performance.

I was very pleased when my best meters return consistent results with average AC, RMS AC, and RMS AC+DC measurement of my DC calibrated precision square wave.

[mawyatt]

Is it necessary to have a low distortion sine-wave for checking AC meter performance? As mentioned we used a square-wave since producing and accurate squarewave is straightforward using CMOS logic and flip-flops.

Seems that as long as the distortion products don't extend beyond the measuring instrument meter bandwidth, that they get accurately accounted for in the RMS computations, or in the analog RMS converter chip, which ever the meter utilizes.

For a basic check perhaps not, but for calibration to within the meters specs, you probably need a low-THD sine as specified.

One thing to think about is that in addition to any error that might occur simply because of dissimilar RMS conversion of non-sinusoids, the meters have a frequency-dependent accuracy spec.  Higher frequencies result in wider tolerances.  Take your square wave calibrator and look at how much energy there is in those upper harmonics and which tolerance bin they fall into.  Say you have a meter with a specified accuracy of +/- 0.02% @ 1kHz, but +/- 3% @ 50kHz.  What percentage of the signal is in that 49th or 51st harmonic?  How about the percentage of the signal that is in all the harmonics above 50kHz?  And how does the 3% error multiplied by that  percentage compare with the 0.02% error at the fundamental? 

Now if you are talking about two sine stimuli, one with 1% THD and the other with 0.1% THD, both primarily in the 2nd and 3rd harmonics, then perhaps the difference will be less.  This would not be difficult to test.  Perhaps the next time I'm working on a meter or a calibrator, I'll try it.

Agree, one could calculate the effects harmonics have based on the meter characteristics vs frequency and as enut11 mentioned because the square-wave has such high odd order harmonics (squarewave vary as 1/n, triangle is 1/n^2 I recall) this limits the squarewave approach to a fairly low frequency.

Just did a quick set of measurements using one of those cheap XAR chip generators that produce horrible looking sine (and triangle) waves at 1KHz. The 2 year old KS34465A and 2 week old DMM6500 agree very well and use the same computational RMS method I believe. The two 33401As agree with each other, and with the DMM6500 within ~50ppm, but use an analog RMS conversion chip like the Siglent SDM3065X which differed by the highest at ~800ppm. Then changed to a ratty looking triangle wave, with almost same results.

Did same tests with better sine and triangle waveform AWG with good results, but actually the results from the KS34465A, DMM6500 and two 34401As were even more aligned with the "ratty" waveforms from the cheap XAR generator

The Siglent SDM3065X didn't agree well with any of the other meters, regardless of the waveforms used, but does use an analog RMS conversion chip, however so does the two 34401As, so go figure

Best,

[trobbins]

There are a few USB soundcards that have adequate audio bandwidth out to 80-90kHz before amplitude droop becomes noticeable.  The advantage of that type of oscillator source is that software can be used to null out harmonics from the 'output'.  As such, a very low distortion output can be sourced from the soundcard, but also extend further out to include any following buffer amp, as well as any step-up audio transformer.  Nulling harmonics in that manner can achieve low distortion levels that are intrinsic in just the ADC of the soundcard, and of course if the frequency is not too high in that bandwidth (eg. circa 20kHz fundamental for first few harmonics to be nulled), and the same applies for distortion introduced by a step-up transformer at the low-frequency end.  Certainly a cheap and easy way to start testing without the need for a low distortion oscillator.

Using a step-up audio transformer may benefit from some judicious loading of the high voltage output winding to maximise high frequency response flatness.  The output transformer likely needs to be 'hi-fi' in design to both push out amplitude variations from the first resonance, and to constrain the 'Q' of that resonance, given that resonance could well be below 50kHz depending on the transformer (a good transformer can push that resonance out to circa 100kHz).  The soundcard technique with suitable software (like REW) can even test the impedance of the transformer to confirm where the first resonance occurs (and hence the likely limit to frequency response for high voltage calibration efforts).

[mawyatt]

Is it necessary to have a low distortion sine-wave for checking AC meter performance? As mentioned we used a square-wave since producing and accurate squarewave is straightforward using CMOS logic and flip-flops.

Seems that as long as the distortion products don't extend beyond the measuring instrument meter bandwidth, that they get accurately accounted for in the RMS computations, or in the analog RMS converter chip, which ever the meter utilizes.

Might expect some differences with meters that use different RMS methods, like computational or analog, but not so much between "like" technique instruments, again as long as the BW isn't pushed.

I don't think DMMs are generally specified for square(ish) wave signals. I would definitely want to use a bandwidth-limited signal limited well below the bandwidth of all meters involved. Then I imagine would kind of work, but I'd expect the uncertainty of your comparison to be higher than with sine waves due to the different frequencies involved and the finite attenuation of any filter.

Performance verification generally calls for frequencies across the bandwidth. For example for the HPAK 34401A, they use frequencies ranging from 20 Hz to 300 kHz (there's a reason why the Fluke 5200A has such a wide bandwidth). How would you verify the high end. A 300 kHz square wave? A 100 kHz square wave limited to 300 kHz (won't be very square)?

This technique is simply a method to allow an accurately known voltage reference to the translated to an accurate RMS waveform within limits. Since a perfect squarewave has an RMS value (by definition) of the peak value. Same goes for a DC value, RMS is (by definition) equal to the DC level, and the average of unipolar squarewave is 1/2 the the peak value. A low frequency CMOS output from a Flip-Flop is almost a perfect squarewave with 50% duty cycle and swings unloaded from zero the VDD.

So in the AC cal reference mentioned we had an output that was accurate in both RMS and average DC level, and directly related to the 5.000V DC reference utilized for VDD for just a few $, not a bad cost/performance ratio IMO. Would really like a cal standard like the 5200A, but well out of our budget range, so we did the best we could.

Many years ago we developed a precision DC voltage divider that won EDNs Best Idea of the Year (patent 5030848) based upon a similar concept except this patented DC voltage divider had almost no ripple and achieve sub-ppm levels of voltage division accuracy (1/2) without any precision components! The general idea was to use a CMOS FF Q and Qbar output to toggle two resistors connected together, say R1 and R2, and the junction between them is Vout. When Q is high then the output is VDD*(R2/(R1 + R2)), when Q is low the output is VDD*(R1/(R1 + R2)), the average is VDD/2 independent of either R1 or R2. A mismatch in R1 and R2 just causes a small ripple which is filtered with a single shunt C and the output. Of course the CMOS FF output NMOS and PMOS fets are not equal, but can be made insignificant with just a simple buffer. We also did this with a discrete NMOS and PMOS inverter in our AC calibrator.

Anyway, CMOS digital logic can be utilized in many unusual useful ways, these are just a few.

Best,

[enut11]

There are a few USB soundcards that have adequate audio bandwidth out to 80-90kHz before amplitude droop becomes noticeable.  The advantage of that type of oscillator source is that software can be used to null out harmonics from the 'output'.  As such, a very low distortion output can be sourced from the soundcard, but also extend further out to include any following buffer amp, as well as any step-up audio transformer.  Nulling harmonics in that manner can achieve low distortion levels that are intrinsic in just the ADC of the soundcard, and of course if the frequency is not too high in that bandwidth (eg. circa 20kHz fundamental for first few harmonics to be nulled), and the same applies for distortion introduced by a step-up transformer at the low-frequency end.  Certainly a cheap and easy way to start testing without the need for a low distortion oscillator.

Using a step-up audio transformer may benefit from some judicious loading of the high voltage output winding to maximise high frequency response flatness.  The output transformer likely needs to be 'hi-fi' in design to both push out amplitude variations from the first resonance, and to constrain the 'Q' of that resonance, given that resonance could well be below 50kHz depending on the transformer (a good transformer can push that resonance out to circa 100kHz).  The soundcard technique with suitable software (like REW) can even test the impedance of the transformer to confirm where the first resonance occurs (and hence the likely limit to frequency response for high voltage calibration efforts).

Hi @trobbins.
I measured the audio transformer output impedance at around 1,700 ohms at 1KHz using a resistive load. Can you suggest what sort of transformer loading I could experiment with to maximise high frequency response?
Also, at the moment, the amp output is loaded only by what it sees at the 8ohm audio transformer winding.

[trobbins]

Perhaps start with an 8k resistive load that reflects 8 ohm back to the amplifier (eg. for the M1120, the 1.25W tap presents 8kohm for an 8 ohm speaker load, with a 31.6:1 turns ratio).  You could sweep through the frequency range to see what voltage response you get, but at low voltage (as an 8k load would consume some power for higher voltages).  You may still get a reasonably flat response for higher resistance loads.

[ledtester]

TheHWCave did a couple of videos on building a power amplifier for a function generator based on a OPA541 module available from aliexpress...

Building an OPA541-based power amplifier for function generators
https://youtu.be/925X0RDUy9w

#103: Some changes to my OPA541-based power amplifier for function generators
https://youtu.be/ERJ9Om-eBqo

It was used to test the accuracy of a PZEM-004T AC power meter:

How accurate is the Peacefair PZEM-004T AC Comms Module?
https://youtu.be/j0_y8dPfpKc

The setup in that case was: FY6600 -- Power Amp -- 5V/240V transformer

[alm]

This technique is simply a method to allow an accurately known voltage reference to the translated to an accurate RMS waveform within limits. Since a perfect squarewave has an RMS value (by definition) of the peak value. Same goes for a DC value, RMS is (by definition) equal to the DC level, and the average of unipolar squarewave is 1/2 the the peak value. A low frequency CMOS output from a Flip-Flop is almost a perfect squarewave with 50% duty cycle and swings unloaded from zero the VDD.

The Tektronix PG506 calibration generator (for scope calibration) used the same technique of providing both the DC and chopped DC level, although for oscilloscopes, so the accuracy was only 0.25%. The problem for DMMs is the probably imperfect response of the DMM to this almost perfect squarewave.

[mzzj]

There are a few USB soundcards that have adequate audio bandwidth out to 80-90kHz before amplitude droop becomes noticeable.  The advantage of that type of oscillator source is that software can be used to null out harmonics from the 'output'.  As such, a very low distortion output can be sourced from the soundcard, but also extend further out to include any following buffer amp, as well as any step-up audio transformer.  Nulling harmonics in that manner can achieve low distortion levels that are intrinsic in just the ADC of the soundcard, and of course if the frequency is not too high in that bandwidth (eg. circa 20kHz fundamental for first few harmonics to be nulled), and the same applies for distortion introduced by a step-up transformer at the low-frequency end.  Certainly a cheap and easy way to start testing without the need for a low distortion oscillator.

Using a step-up audio transformer may benefit from some judicious loading of the high voltage output winding to maximise high frequency response flatness.  The output transformer likely needs to be 'hi-fi' in design to both push out amplitude variations from the first resonance, and to constrain the 'Q' of that resonance, given that resonance could well be below 50kHz depending on the transformer (a good transformer can push that resonance out to circa 100kHz).  The soundcard technique with suitable software (like REW) can even test the impedance of the transformer to confirm where the first resonance occurs (and hence the likely limit to frequency response for high voltage calibration efforts).

I have been thinking of similar harmonic cancellation. Is there a ready plug-and-play software already available for that?

Having said that I think it is huge overkill for DMM calibration. Even best calibrators like Fluke 5730A specify distortion at something like 0.03%
Any decent audio gear is easily in -90dB level or 0.003%.

---------
ESL (electrostatic loudspeaker) folks might have some clever ideas on how to get some voltage swing without transformer coupling the output.
Full-bridge output with cascoded 1kV mosfets would give you easily  over 1kVrms output. 
 

[trobbins]

mzzj, Room Equalise Wizard (REW) has it in their generator.  The spectrum analyser is used to view the 'output' signal to be nulled and each harmonic's amplitude and phase can be adjusted to null that harmonic down in to the noise floor, with settings for up to 10th harmonic.  The null degrades as frequency and amplitude are changed, as expected.  It's just convenient if you have a need.

[tszaboo]

For generating AC signals for a microcontroller, or something like that, the easiest way is with DDS generators from AD. Select one that has a few MHz range only. You can feed this into an amplifier for gain, and a second one for low impedance output if needed. If you want to measure AC (for feedback for example) similarly, probably the easiest to do is with a RMS to DC converter chip from AD. The LT parts from their portfolio are using delta-sigma modulation to do this conversion, and they are quite accurate once calibrated.

[enut11]

The prototype variable voltage/frequency AC Calibrator is now working to my satisfaction. Provided enough time is given for the setup to settle wrt temperature (>60min) the stability is good enough to allow a valid simultaneous comparison of two or more meters. Maximum frequency is about 50KHz for LV out and 15KHz for HV out.

This design should only be used with pure sine wave input. Square waves can give unpredictable results at low frequencies (core saturation) and high frequencies (attenuation).

I have de-rated the maximum AC outputs to 10vACrms (LV) and 100vACrms (HV). The latter was purely a choice based on the 100v rating of the Jaycar MM1900 5W audio output transformer.

I added a 1Kohm input voltage trim 10-turn pot. This sits at the top of the amplifier 100Kohm input resistor and gives a fine adjustment of about 1% of input signal and allows better than 1mV resolution at 100vAC signal out.

There are still parts of the build that could be improved but that will be part of phase 2 of this project.

[David Hess]

If you want to measure AC (for feedback for example) similarly, probably the easiest to do is with a RMS to DC converter chip from AD. The LT parts from their portfolio are using delta-sigma modulation to do this conversion, and they are quite accurate once calibrated.

If I wanted to measure RMS with as much accuracy as possible, then I would use a high resolution sampling converter and calculate it directly.  Such a sampling converter could be integrated, or if I thought I could get better performance, I might use a discrete high precision sampler in front of a slower but more accurate converter.

The most significant limitation to accuracy would be the passband response of the stages up to and including the sampler.  If the frequency content can be known, then correction could be applied after conversion.

[1audio]

However if you want to be able to compare to the international standards you need to use a thermal converter. They are finicky, fragile and drift a lot but they are still the gold standard for AC.

I was thinking about the limitations of the square wave calibration. You could predict the error from the rolloff I believe and possibly figure out the bandwidth of the meter and potential HF accuracy using several frequencies. Ultimately you would need to compare back to a sine wave reference to be sure. I don't think its a dead end but the bandwidth correction will be essential for accuracy. I'm not sure how something like the HP3458 would react to that, the HP 3458 uses sampling in effect for AC measurement so some strange things could happen at certain frequencies.

The current best efforts for AC at midband (100 Hz to 10 KHz) are around 10 ppm or so.

HP made a sampling AC voltmeter that was good to 1 GHz around 50 years ago. They abandoned it for some reason, like no customers? They also made a Thermal RF power meter with higher accuracy (a version is still in production for $20K +) which does still have customers.

[enut11]

However if you want to be able to compare to the international standards you need to use a thermal converter. They are finicky, fragile and drift a lot but they are still the gold standard for AC.

I was thinking about the limitations of the square wave calibration. You could predict the error from the rolloff I believe and possibly figure out the bandwidth of the meter and potential HF accuracy using several frequencies. Ultimately you would need to compare back to a sine wave reference to be sure. I don't think its a dead end but the bandwidth correction will be essential for accuracy. I'm not sure how something like the HP3458 would react to that, the HP 3458 uses sampling in effect for AC measurement so some strange things could happen at certain frequencies.

The current best efforts for AC at midband (100 Hz to 10 KHz) are around 10 ppm or so.

HP made a sampling AC voltmeter that was good to 1 GHz around 50 years ago. They abandoned it for some reason, like no customers? They also made a Thermal RF power meter with higher accuracy (a version is still in production for $20K +) which does still have customers.

Hi @1audio. Your are correct of course with your comments about the thermal converter method being the gold standard. However, it is a finicky method and requires systematic best practice to work well. I wanted a 'quick and dirty' method for comparative measurements. With my setup, assuming you have a good reference meter, you can compare several instruments simultaneously under arguably identical conditions.
enut11

[tszaboo]

If you want to measure AC (for feedback for example) similarly, probably the easiest to do is with a RMS to DC converter chip from AD. The LT parts from their portfolio are using delta-sigma modulation to do this conversion, and they are quite accurate once calibrated.

If I wanted to measure RMS with as much accuracy as possible, then I would use a high resolution sampling converter and calculate it directly.  Such a sampling converter could be integrated, or if I thought I could get better performance, I might use a discrete high precision sampler in front of a slower but more accurate converter.

The most significant limitation to accuracy would be the passband response of the stages up to and including the sampler.  If the frequency content can be known, then correction could be applied after conversion.

I think you underestimate the complexity of such systems.
Let's take the LTC1968. It will work to 500khz without too much trouble, costs 4 USD and it needs one capacitor (film preferably) and a slow 12-16 bit ADC to digitize it.
If you want to make a similar system with sampling, you have to make a DC ad AC accurate antialiasing filter, Sample it with an ADC at least 1MSPS, but realistically, much higher, I would say 5MSPS. Then you have to make sure that your sampling frequency is not the multiple of the incoming signal, because then you sample it the same phase. Or you undersample it, and use an ADC with high input bandwidth. It will be a SAR ADC in either case, others will not have the sample rate.
Nevertheless, driving these is not trivial, needs a high bandwidth opamp, in the order of hundreds of MHz otherwise the input of the ADC will miss codes and all kinds of nastiness.
Then we arrive at the microcontroller.
When you all verified the system, wrote the code, and finished it up... Sounds like a lot of work doesn't it? Could be more accurate? Yes, sure, but be ready to those debugging sessions, when "I wish I would have a 16 bit scope to see what my ADC driver is doing, and why do I have INL errors in my output code".

[1audio]

And you would still need one of those infernal thermal voltage converters to validate your efforts. The whole thing is a rabbit (rat) hole. Just sorting out my vintage calibrator has been a challenging exercise in who do you trust? Every instrument deviates as the frequency goes up and not in the same ways.

[David Hess]

HP made a sampling AC voltmeter that was good to 1 GHz around 50 years ago. They abandoned it for some reason, like no customers? They also made a Thermal RF power meter with higher accuracy (a version is still in production for $20K +) which does still have customers.

Racal-Dana had one also.  I have a couple of Tektronix 7S11 samplers so could put something with similar or better performance together in about 10 minutes with what I have on hand.  Tektronix even published an application note showing how to do it.

The advantage of sampling measurement over linear measurement (for lack of a better term) is that the frequency response of the sampler can be calculated from the sampling gate width which can be measured with an unleveled RF source.  So if you know the frequency components of the measured signal, which will often be the case with an RF sampling voltmeter measurement, a simple lookup will correct for the passband response.  Racal-Dana included the correction curve for their sampling RMS voltmeter in their manual.

If I wanted to measure RMS with as much accuracy as possible, then I would use a high resolution sampling converter and calculate it directly.  Such a sampling converter could be integrated, or if I thought I could get better performance, I might use a discrete high precision sampler in front of a slower but more accurate converter.

The most significant limitation to accuracy would be the passband response of the stages up to and including the sampler.  If the frequency content can be known, then correction could be applied after conversion.

I think you underestimate the complexity of such systems.
Let's take the LTC1968. It will work to 500khz without too much trouble, costs 4 USD and it needs one capacitor (film preferably) and a slow 12-16 bit ADC to digitize it.

If you want to make a similar system with sampling, you have to make a DC ad AC accurate antialiasing filter,

Nope, one of the chief advantages of the sampling RMS measurement is that no antialiasing is required at all.

But what I was suggesting is either precision sampling followed by slow precision RMS conversion, or the direct computational approach where a high frequency high resolution sampling ADC is used for low frequency RMS conversion.  I was told about 20 years ago that some multimeters had started using the later method in lieu of analog computation but have not been able to verify that.

The Racal-Dana and HP RF voltmeters mentioned above use the former method, and I am suggesting extending this idea to low frequency measurement.

Sample it with an ADC at least 1MSPS, but realistically, much higher, I would say 5MSPS. Then you have to make sure that your sampling frequency is not the multiple of the incoming signal, because then you sample it the same phase. Or you undersample it, and use an ADC with high input bandwidth. It will be a SAR ADC in either case, others will not have the sample rate.

The bandwidth is determined by the sampling gate width, which produces a non-linear passband response that can be easily calculated.

I suggested a simple system relying on a sampling ADC with high input bandwidth used at much lower frequencies, but an alternative is to implement a precision external sampler with a slower ADC.  I messed around with the later many many years ago but not with this application in mind and I mostly learned what does not work for precision sampling.

Nevertheless, driving these is not trivial, needs a high bandwidth opamp, in the order of hundreds of MHz otherwise the input of the ADC will miss codes and all kinds of nastiness.

The chief advantage of the sampling method is that within the input signal range, no processing is required before sampling, removing those error contributions, which is the major reason to use it.  I can design and fabricate  a sampler or sampling system, but constructing a high accuracy thermal RMS converter would be much more difficult, at least for me.

Then we arrive at the microcontroller.
When you all verified the system, wrote the code, and finished it up... Sounds like a lot of work doesn't it? Could be more accurate? Yes, sure, but be ready to those debugging sessions, when "I wish I would have a 16 bit scope to see what my ADC driver is doing, and why do I have INL errors in my output code".

Calibration requires an unleveled RF source to measure the sampling gate width.  If you are desperate, processing after sampling can be done with any low frequency RMS measuring tool.  I could set something up which makes RMS measurements to GHz frequencies in about 10 minutes with what I have on hand, but it would not be real suitable for the low frequency high resolution measurements we are discussing, although it would work.

[tszaboo]

HP made a sampling AC voltmeter that was good to 1 GHz around 50 years ago. They abandoned it for some reason, like no customers? They also made a Thermal RF power meter with higher accuracy (a version is still in production for $20K +) which does still have customers.

Racal-Dana had one also.  I have a couple of Tektronix 7S11 samplers so could put something with similar or better performance together in about 10 minutes with what I have on hand.  Tektronix even published an application note showing how to do it.

The advantage of sampling measurement over linear measurement (for lack of a better term) is that the frequency response of the sampler can be calculated from the sampling gate width which can be measured with an unleveled RF source.  So if you know the frequency components of the measured signal, which will often be the case with an RF sampling voltmeter measurement, a simple lookup will correct for the passband response.  Racal-Dana included the correction curve for their sampling RMS voltmeter in their manual.

If I wanted to measure RMS with as much accuracy as possible, then I would use a high resolution sampling converter and calculate it directly.  Such a sampling converter could be integrated, or if I thought I could get better performance, I might use a discrete high precision sampler in front of a slower but more accurate converter.

The most significant limitation to accuracy would be the passband response of the stages up to and including the sampler.  If the frequency content can be known, then correction could be applied after conversion.

I think you underestimate the complexity of such systems.
Let's take the LTC1968. It will work to 500khz without too much trouble, costs 4 USD and it needs one capacitor (film preferably) and a slow 12-16 bit ADC to digitize it.

If you want to make a similar system with sampling, you have to make a DC ad AC accurate antialiasing filter,

Nope, one of the chief advantages of the sampling RMS measurement is that no antialiasing is required at all.

But what I was suggesting is either precision sampling followed by slow precision RMS conversion, or the direct computational approach where a high frequency high resolution sampling ADC is used for low frequency RMS conversion.  I was told about 20 years ago that some multimeters had started using the later method in lieu of analog computation but have not been able to verify that.

The Racal-Dana and HP RF voltmeters mentioned above use the former method, and I am suggesting extending this idea to low frequency measurement.

Sample it with an ADC at least 1MSPS, but realistically, much higher, I would say 5MSPS. Then you have to make sure that your sampling frequency is not the multiple of the incoming signal, because then you sample it the same phase. Or you undersample it, and use an ADC with high input bandwidth. It will be a SAR ADC in either case, others will not have the sample rate.

The bandwidth is determined by the sampling gate width, which produces a non-linear passband response that can be easily calculated.

I suggested a simple system relying on a sampling ADC with high input bandwidth used at much lower frequencies, but an alternative is to implement a precision external sampler with a slower ADC.  I messed around with the later many many years ago but not with this application in mind and I mostly learned what does not work for precision sampling.

Nevertheless, driving these is not trivial, needs a high bandwidth opamp, in the order of hundreds of MHz otherwise the input of the ADC will miss codes and all kinds of nastiness.

The chief advantage of the sampling method is that within the input signal range, no processing is required before sampling, removing those error contributions, which is the major reason to use it.  I can design and fabricate  a sampler or sampling system, but constructing a high accuracy thermal RMS converter would be much more difficult, at least for me.

Then we arrive at the microcontroller.
When you all verified the system, wrote the code, and finished it up... Sounds like a lot of work doesn't it? Could be more accurate? Yes, sure, but be ready to those debugging sessions, when "I wish I would have a 16 bit scope to see what my ADC driver is doing, and why do I have INL errors in my output code".

Calibration requires an unleveled RF source to measure the sampling gate width.  If you are desperate, processing after sampling can be done with any low frequency RMS measuring tool.  I could set something up which makes RMS measurements to GHz frequencies in about 10 minutes with what I have on hand, but it would not be real suitable for the low frequency high resolution measurements we are discussing, although it would work.

OK, an RF sampling ADC. I'm not familiar with those, so actually don't know how well they would work, so I take your word for it.
I can tell you that this wouldn't work with a SAR ADC.
Those LTC196x chips actually are not thermal based, they are using some sigma-delta conversion, and they are super simple to use. But I guess, each to their own.

[David Hess]

OK, an RF sampling ADC. I'm not familiar with those, so actually don't know how well they would work, so I take your word for it.

I can tell you that this wouldn't work with a SAR ADC.

It would work with a *sampling* SAR ADC (not all SAR ADCs are sampling ADCs), and it is what I would try first, but performance will be limited by the relatively low input bandwidth.  Some modern parts are very fast now.

I have been told that some modern multimeters do exactly this with a fast high resolution sampling SAR ADC operating at a relatively low sample rate and RMS conversion done on the digital side, but I have not confirmed it directly.

Those LTC196x chips actually are not thermal based, they are using some sigma-delta conversion, and they are super simple to use. But I guess, each to their own.

I suspect they use switched capacitor computation techniques.

[Kleinstein]

I have been told that some modern multimeters do exactly this with a fast high resolution sampling SAR ADC operating at a relatively low sample rate and RMS conversion done on the digital side, but I have not confirmed it directly.

The relatively popular DMM chip set HY3131 (e.g. used in one of Daves DMMs) have the option to use digital RMS with a relatively fast internal SD ADC.
The BW is still a bit limited in this case, and an external RMS chip may still be used. Both ways have there advantages.
The new DMM6500 is supposed to use digital RMS with an extra ADC chip (should be SAR type). Chances are they added the fast 2 nd ADC for the digital RMS and than also used it for the ditizing mode.

The  LTC196x would use switched capacitors like other SD ADC chips and many modern SAR ADC chips. So the driving needs can be similar.

For the RMS measurement a defined bandwidth can be an advantage. Higher BW is not allways better as this also means more noise. Especially with a calibrator the signal is relatively well behaved with a known frequency and low crest factor. The higher harmonics should not contribute too much. So things are a bit easier than with a normal DMM input. If filtering is needed some of this could be before the output and no extra anti aliasing needed at the ADC.
For the RMS calculation alising can be good and include frequencies higher than Fs without much extra effort. So the limit is not from the sampling rate, but the sampling part of the ADC and the input amplifier.

[enut11]

All interesting stufff guys but what I would like at the moment is feedback on the analog building block solution that I am working on.
My approach has been to use better components. Is there any practical way to improve the design?
How can the 3 major components, sig gen, power amp and output transformer, all currently independent, be coupled into an effective AC Voltage calibrator?
enut11

[mawyatt]

Not sure exactly how the DMM6500 performs RMS, but it agrees very well with the KS34465A which also uses a digital computational method. Maybe the Keysight and Keithley methods are similar. We found the old 34401A using the analog method also agree well with the DMM6500 which is our latest DMM only a few weeks old. Superb instrument BTW!!

I guess one could assume that both methods should give acceptable results if the Crest Factor and Waveform Fundamental Frequency are low enough. That was the basis for using the squarewave as a comparison waveform, since it's relatively easy to get a precise known peak value.

Best,

[2N3055]

Not sure exactly how the DMM6500 performs RMS, but it agrees very well with the KS34465A which also uses a digital computational method. Maybe the Keysight and Keithley methods are similar. We found the old 34401A using the analog method also agree well with the DMM6500 which is our latest DMM only a few weeks old. Superb instrument BTW!!

I guess one could assume that both methods should give acceptable results if the Crest Factor and Waveform Fundamental Frequency are low enough. That was the basis for using the squarewave as a comparison waveform, since it's relatively easy to get a precise known peak value.

Best,

There is a Keithley engineer dropping by on the forum occasionally. Real nice guy, too..
RMS method was mentioned and he said DMM6500 uses sampling + RMS calculation. No exact details of implementation, but sampling it is. There is no RMS converter chip inside...

[David Hess]

I have been told that some modern multimeters do exactly this with a fast high resolution sampling SAR ADC operating at a relatively low sample rate and RMS conversion done on the digital side, but I have not confirmed it directly.

The relatively popular DMM chip set HY3131 (e.g. used in one of Daves DMMs) have the option to use digital RMS with a relatively fast internal SD ADC.
The BW is still a bit limited in this case, and an external RMS chip may still be used. Both ways have there advantages.

That might explain what I was told and why I could not confirm it.  In a discussion about the Fluke lawsuit, it was reported to me that the Tektronix DMM916 did exactly what I described, but inspection of the printed circuit board shows an Analog Devices AD737, and the converter chip datasheet reveals no capability for digital RMS processing.  Of course the report may have been referring to something external to both.

The  LTC196x would use switched capacitors like other SD ADC chips and many modern SAR ADC chips. So the driving needs can be similar.

And like Linear Technologies long experience with switched capacitor filters.

For the RMS measurement a defined bandwidth can be an advantage. Higher BW is not allways better as this also means more noise. Especially with a calibrator the signal is relatively well behaved with a known frequency and low crest factor. The higher harmonics should not contribute too much. So things are a bit easier than with a normal DMM input. If filtering is needed some of this could be before the output and no extra anti aliasing needed at the ADC.

For the RMS calculation alising can be good and include frequencies higher than Fs without much extra effort. So the limit is not from the sampling rate, but the sampling part of the ADC and the input amplifier.

The tradeoff is that the sampling bandwidth, and its noise contribution, is fixed by the sampling gate width and no filtering before can reduce that, and no filtering after it can remove aliased noise.  So input bandwidth to some arbitrary cutoff can be extended, but only at the expense of increased noise unless the sampling rate is also increased to prevent aliasing.

I seem to recall that about 50 kHz was the sweet for sampling rate in old designs to allow reasonable operating life of the needed avalanche pulse generator.  A precision sampler would work differently of course.

[enut11]

Perhaps start with an 8k resistive load that reflects 8 ohm back to the amplifier (eg. for the M1120, the 1.25W tap presents 8kohm for an 8 ohm speaker load, with a 31.6:1 turns ratio).  You could sweep through the frequency range to see what voltage response you get, but at low voltage (as an 8k load would consume some power for higher voltages).  You may still get a reasonably flat response for higher resistance loads.

Thanks @trobbins. I tried an 8.2K 2W resistor across the transformer high voltage output and it had a big affect in flattening the high frequency response. Good suggestion
enut11

[Kleinstein]

The signal genrator is relatively straight forward these days.  DDS genration should give a pretty stable and clean enough sine wave. Using a wien bridge or similar can get lower distortion but much more tricky to stabilize the frequency and amplitude.
The power amplifier part is the easy part. At up to some 20 kHz audio should be good enough. The tricky part could be however if the transformer is to be driven in closed loop and not open loop.
I an afraid that a more normal transformer may not be very linear. It depends quite a bit on the type and amplitude to use. A simpel resistive load should not make that much difference. So ideally one would need some feedback directly from the output. One may get away with direct linear FB to have the transformer in the ampliers loop. The alternative may be feedback via a way to measure the amplitude. This could be true RMS, but also simple rectification, as the waveform is known. With a measurement  fron the output, the stability of the generator, amplifer and transformer would no longer be important.

In the extremes there would be the option to have a special precision transformer with seprate drive, correct and sense. This can make up a very stable and accurate gain, but at high costs / effort.

[enut11]

I have carried out some spot frequency checks on the Jaycar MM1900 5W audio transformer (HV Out). Response is flat up to ~30KHz and useful to around 50KHz. The higher frequencies improved dramatically after loading the transformer output with an 8.2Kohm 2W resistor.

Remember, the AC Voltage Calibrator is just a stable AC voltage/frequency source relying on a known reference meter to measure the actual value and transfer this to another instrument.
enut11

[enut11]

Frequency response of the LM1875 power amp was also checked and found to be good to around 75KHz.

I had to change the recommended filter on the LM1875 amp output (1ohm+0.22uF) to prevent the resistor from overheating at high frequencies.  I used a 15ohm 1W resistor in series with a 0.022uF cap.

Ti also recommend the following for high frequency stability:
"A method commonly employed to protect amplifiers from low impedances at high frequencies is to couple to the load through a 10Ω resistor in parallel with a 5 μH inductor."

[Kleinstein]

The RC daming element at the amplifier output is there for a reason. Changing this comes with the risk of oscillation or ringing. So one should at least check with the scope and square wave signal if the ringing gets excessive.

The series RL element can help has it isolates possible capacitive load. It comes with the downside of possibly a slight faster drop in the amplitude for higher frequencies. So one may have to use a smaller inductance. In audio ampliers the inductor is often just some 5 turns around the resistor so more like 0.2 µH.

[enut11]

Thanks @Kleinstein.  For the amplifier only (LVout), I tried both a 1KHz and 10KHz square wave at full output voltage (30vPP) and both were clean. Is this a good enough test for HF stability?

[Kleinstein]

A square wave from the input side is a reasonable test, at least for the lower frequency part. It however does not really test possible ringing mich higher than the amplifiers BW. It can be enough for the test, but it may miss the upper frequency end, which is often the point where ringing happens. The case with no loading to the amplifier is also the easy case and should be OK. It may get tricky with some added capacitive load, even of just some 1 nF (e.g. 10 m of coax). Ringing gets sometimes stronger with smaller amplitude, so that the amplifier is not at it's slew rate limit. The frequency does not really matter, it is about the edges.

[bdunham7]

It however does not really test possible ringing mich higher than the amplifiers BW. It can be enough for the test, but it may miss the upper frequency end, which is often the point where ringing happens.

Just to reinforce that point, when a 'typical' Class AB audio amplifier starts oscillating do to a malfunction, design issue or replacement part difference, the frequency of the oscillation is often several MHz.  So depending on the exact amplifier involved and what sort of input response it has, I suppose ringing might be up there as well.

[trobbins]

enut11, just to clarify can you confirm your scope has sufficiently high bandwidth for the square wave test, and the squarewave input has sufficiently low rise time, and what the load (or loads) were, and whether you had added any load RC or RL networks?

[enut11]

Thanks @bdunham7. I did notice a little VHF noise dancing on top of the 30v square square wave. Don't know if this is normal for an audio amp or what I can do about it though.

[enut11]

Hi @trobbins. My scope is a 100MHz Tekway. The 10KHz 30v squarewave output (LVout) is shown below. The 20KHz test was little different.

[trobbins]

Ta, just for clarity, did the generator output squarewave have a much faster risetime than circa 2.8us (as shown for the power amp LV output signal)?

[enut11]

Wavetek Model 80 function generator wave specs attached:

[TiN]

Just for reference, 3458A can do ACV measurements in multiple ways, with RMS to DC conversion (ACV ANALOG mode), by accurate sampling (ACV SYNC mode) and also by fast sampling (SSDC and SSAC modes). I would highly recommend Rado Lapuh's book showcasing all the various operation modes and benchmarks of 3458A in terms of AC voltage.

There is also very nice appnote showing methods to measure ACV to very high precision from designer of 3458A ACV function, Robert Swerlein.

[enut11]

Hi @TiN. It is interesting to see that the HP3458A is fast enough to be able to accurately sample an AC waveform in DC mode. Amazing.

[1audio]

Setting up an amp with an output transformer works OK. Ideally feedback around the transformer would help. Even if its just a leveling loop since the transformer/amp will have a rolloff and will be sensitive to the load. I have done that in the past with LM3886's and transformers to make a 120V AC supply. Stability is a little tricky but worth the effort.

Is there any info on the amplitude stability of the DDS sources? That stability is critical to the utility of the project. If the voltage is changing (like the 1/2 hour warmup) then it can all be very frustrating.

[David Hess]

Is there any info on the amplitude stability of the DDS sources? That stability is critical to the utility of the project. If the voltage is changing (like the 1/2 hour warmup) then it can all be very frustrating.

Stability of the analog output from a DDS depends on the reference, digital-to-analog converter, and signal conditioning circuitry between the DAC and output.  They are not usually designed for DC precision, but could be.

Hi @TiN. It is interesting to see that the HP3458A is fast enough to be able to accurately sample an AC waveform in DC mode. Amazing.

I have used many different high resolution bench multimeters with triggering capability to make low frequency AC measurements in DC mode.  The low frequency limit depends on how long a time period the measurements are made, and the high frequency limit usually depends on the integration time which is typically an integer number of power line cycles, but can be much shorter at lower resolution.  This is a great way to make 0.1 to 10 Hz noise measurements.

I used this method to design some incredibly low noise DC amplifiers.

[TiN]

I've got ManateeMafia's Holt HCS-1AF on a bench as well and can do some measurements with them. If anyone have particular interest, feel free to drop test suggestions. Sadly 20A shunt from a set is blown open, but other seem to be ok (5A, 2A, 0.5A, 0.2A and 0.02A).

[enut11]

Setting up an amp with an output transformer works OK. Ideally feedback around the transformer would help. Even if its just a leveling loop since the transformer/amp will have a rolloff and will be sensitive to the load. I have done that in the past with LM3886's and transformers to make a 120V AC supply. Stability is a little tricky but worth the effort.

Is there any info on the amplitude stability of the DDS sources? That stability is critical to the utility of the project. If the voltage is changing (like the 1/2 hour warmup) then it can all be very frustrating.

Hi @1audio. Do you have a circuit diagram to illustrate your point? At the moment, the audio transformer output in this project is fully floating.

[enut11]

I have been gathering parts for an upgraded AC Voltage Calibrator, including a larger box. Picture shows the main components to be incorporated. Still waiting for the toroidal power transformer with two 25v windings. This will extend the calibrator low voltage range to just over 20v RMS.

The LV  output will be 0-20v RMS, 20Hz-50KHz (direct power amp output)
The HV output will be up to 200v RMS and 50Hz-25KHz (power amp + audio transformer)

I have a HUA 0-300v AC analog voltmeter which was to go into my 50Hz variac. I tested the meter frequency response and was surprised that it still registered 95% of the true value up to about 50KHz! This will now be used to give a rough indication of voltage on the output terminals. Important from a safety point of view when dealing with voltages over 50vAC. I am also thinking of other ways to indicate the presence of dangerous voltages.

[enut11]

This is the new AC voltage amplifier module. Based on the Hummingbird audio amp design from Silicon Chip (Aust) magazine December 2021 - see reply #14. The only change I made was  smaller, lower power output transistors (TIP41C and TIP42C).

The first question that may come to mind is, why such a big heatsink? The short answer is to control heat - the nemesis of all precision instruments. Class AB low distortion amplifiers draw a significant standing current. In this case 50mA, and this generates approximately 3 watts with no output signal. The 12v fan will be run at 8v to reduce noise.

The module has a gain of ~18 and it will amplify the audio oscillator signal  to 0-20vRMS for frequencies 10Hz-50KHz. This should be suitable for calibrating the lower AC voltage ranges of most multimeters.

[Kleinstein]

With smaller power transistors, the emitter resistors should also be larger. This makes the adjustment of the standing current more stable and may allow to run the amplifier with less current.  A little more current flow still helps to make the amplifier fast. So some power loss is normal and with an instrument not used 24/7 should not be so bad.
If speed of the amplifier gets a problem, I would consider the slightly faster D44H / D45H instead of the TIP41/42.

[enut11]

Thanks @Kleinstein. I have replace the output resistors (0.22ohm) with 1 ohm 1W and reset the bias current to ~25mA, a little lower than spec but the waveform is clean and the amp runs much cooler. Tested to 15vRMS running off a power supply. The full output of 20vRMS will have to wait until the power transformer arrives.
enut11

[1audio]

RE feedback around the transformer. You want to lower the output impedance so that you can be confident in the measured output voltage not changing once set.  Here is feedback around the XFMR. I set up around 6-10 dB of feedback so stability is not really hard to achieve. I have used this to drive a turntable motor with success. I was using LM3886 amp chips which are easy to work with.

The other option, leveling, is more involved.

[Kleinstein]

For the feedback around the transformer the very high frequency part should usually bypass the transformer. This makes it easier to get stability. Obviously the DC part also needs to be from before the transformer and the DC offset may need some care.

[enut11]

For the feedback around the transformer the very high frequency part should usually bypass the transformer. This makes it easier to get stability. Obviously the DC part also needs to be from before the transformer and the DC offset may need some care.

More information please or where I can find some. Working with signal transformers is new to me.
enut11

[1audio]

Unfortunately transformer coupled linear audio amps without tubes are reare so there is not a lot to refer to. Here is some that can help:
https://www.jensen-transformers.com/wp-content/uploads/2014/08/Audio-Transformers-Chapter.pdf
https://www.jensen-transformers.com/wp-content/uploads/2014/08/an001.pdf
https://www.lundahltransformers.com/wp-content/uploads/datasheets/feedbck.pdf     Interesting but tricky to implement. Audio precision has something similar. Details are in their patent: https://patents.google.com/patent/US4614914A/en

You can look at the tube stuff but its not all that detailed in how to stabilize, except for stuff from the 1950s and 1960s.
Using HF feedback to reduce the gain around the transformer is important for stability. Cap coupling to keep DC out may also be important. Even a little DC can saturate the transformer.

[trobbins]

The soundcard/software technique can measure HV output magnitude/phase response before feedback is applied to the output of the step-up transformer, to get an 'open-loop' response to guide whether the HF response is smooth or prone to resonance that may limit or make closed loop operation unstable when applying too much feedback, especially if your HV loading varies a bit.  Or you can do some spot checks with a scope and use X-Y for phase assessment.  As indicated, the feedback signal from after the transformer needs to be rolled-off to reduce HF content level that may cause unstable operation (ie. phase and gain margins need to be kept to respectable levels and not go too close to unstable conditions).

[enut11]

You can look at the tube stuff but its not all that detailed in how to stabilize, except for stuff from the 1950s and 1960s.
Using HF feedback to reduce the gain around the transformer is important for stability. Cap coupling to keep DC out may also be important. Even a little DC can saturate the transformer.

The new Hummingbird amp module I am working with has ~60mV DC at the output. Is this significant wrt transformer saturation or not?

[trobbins]

That is 60mVdc across an 8 ohm winding with a 5W rating that could be used for up to 6.3Vrms across it (ie. 9Vpk).  So no, 60mVdc is not going to bias the core much at all.

[1audio]

The dcr of an 8 Ohm winding is more like .5 Ohms so the current could be significant. Is there an offset trim on the amp?

[enut11]

No output offset is provided on the Hummingbird amp. See reply #63

[1audio]

A basic fix would be a pot across the emitters of the input pair tied to the input current source. 1K should be enough to get adequate adjustment.

[enut11]

The power transformer has arrived and I can now get into part 2 of the AC Calibrator project.

Aim: 0-20vRMS directly out of the power amp 'LV' module for frequencies from 50Hz up to around 50KHz.

This will require a little over 60Vpp or a minimum +/- 30v rails for the power amp. As I intend to regulate the power rails for additional stability, I chose a power transformer with two 25v secondary windings. Rectified and filtered these will give about +/-36v DC.

The voltage regulator kit I used had LM317/LM337 chips. I had to change the 5K pots as they did not allow enough adjustment. I set the regulated outputs to +/- 31v DC. This allowed about 5v (36-31) for the regulators to work.

At lower frequencies (<20KHz) there was no problem generating 20vRMS with this setup. However, as frequency was increased, the amp output capacitor to ground (220nF) has lower and lower reactance and presents a heavier load. At around 48KHz the reactance is ~15 ohm and this, along with the 15 ohm output resistor, puts a load of about 30 ohms on the amp output.

The resulting current of 670mA (20v/30 ohms) is enough to drop the 317/337 pair out of regulation with significant ripple breaking through.

Looking for feedback here. How much leeway do I have to fiddle with the 15 ohm resistor and 220nF cap? Looking to a least halve the amp output loading at 50KHz.

[trobbins]

Perhaps push the zobel corner out one decade to 480kHz, given you have a somewhat controlled 'speaker' loading on the amp.

As an aside, many commercial ss 'brute' PA amps sell for peanuts and have flat frequency/phase response out past 50kHz (I use my American DJ Proformer V1000 for output transformer testing as it gets to a moderate Vrms per channel and can be configured as a balanced amp for PP output transformer testing, and is effectively flat response to 90kHz).

[Kleinstein]

The 15 Ohms and 220 nF are part of the compensation to make the amplifier stable, so it would not oscillate, especially not with a capacitove load. The LR combination at the ouput serves a similar purpose. One may be able to get away with less RC load and larger RL seires impedance. How much is actually needed may needs tests with the actual harware. Quite often the RC element was more like a thing added because it helped in the past with no dailed calculation. The transformer usually has some series inductance anyway. So the transformer is often not such a critical load and may get away with less than a normal cross over circuit.

It could also help to look at the amplifier itself to make it better behaved even with capacitive load. A first step would be to get a model for the simulation to see the more critical points. There may be options with different transistors or maybe more current to the VAS stage to speed up the amplifier.
Besides the load, there could also be a slew rate limit that could be a problem.

Usually audio amplifiers don't use extra voltage regulators for there main supply.

[enut11]

Perhaps push the zobel corner out one decade to 480kHz, given you have a somewhat controlled 'speaker' loading on the amp.

As an aside, many commercial ss 'brute' PA amps sell for peanuts and have flat frequency/phase response out past 50kHz (I use my American DJ Proformer V1000 for output transformer testing as it gets to a moderate Vrms per channel and can be configured as a balanced amp for PP output transformer testing, and is effectively flat response to 90kHz).

@trobbins
I interpret this as substituting a 22nF for the 220nF. This would certainly be a lower load, approx 160 ohm total, at 50KHz. Is there a downside to this approach?

@Kleinstein
Agreed, regulating the rails of an audio power is even frowned upon but in this case I am hoping for better amplitude stability. In all probability,  up to 30KHz would be more than enough to test most meters for AC response.

[Kleinstein]

reducing the capacitor in the zobel element from 220 nF to 22 nF could effect the loop stability. It is not sure how much. For a relatively fast amplifier the 22 nF may still be enough. It really depends on the details (e.g. transistor type, layout,...) of the amplifier and the load how much is needed. Some amplifiers get away without that RC element. The suitable size is more a thing to determine from experiment or maybe a careful simulation.

A flat frequency response to 90 kHz does not guarantee the amplifier could do that at full amplitude. There could still be a slew rate limit and the RC element at the output may overheat. It not so rare to find a burned resistor in the zobel element due to some oscillation of either the amplifier itself or some other stage before.

[trobbins]

You may be able to gain some confidence on HF stability performance of just the ss amp by applying a square wave test with the nominal zobel values in to your known load, and then modifying the zobel cap value - if there is no substantial ringing (and any ringing is say a single much higher frequency) then you should be fine given you have a known load scenario (your transformer likely presents a quite low capacitive impedance above its main resonance around 1kHz, up to when the first HF resonance kicks in).

[enut11]

reducing the capacitor in the zobel element from 220 nF to 22 nF could effect the loop stability. It is not sure how much. For a relatively fast amplifier the 22 nF may still be enough. It really depends on the details (e.g. transistor type, layout,...) of the amplifier and the load how much is needed. Some amplifiers get away without that RC element. The suitable size is more a thing to determine from experiment or maybe a careful simulation.

A flat frequency response to 90 kHz does not guarantee the amplifier could do that at full amplitude. There could still be a slew rate limit and the RC element at the output may overheat. It not so rare to find a burned resistor in the zobel element due to some oscillation of either the amplifier itself or some other stage before.

The Humingbird designer quotes amplifier response (-3dB) to 150KHz. I do not know how to test for HF instability. Would this be at rest or with an input signal?
EDIT: I think @trobbins answered my question.

[David Hess]

The Humingbird designer quotes amplifier response (-3dB) to 150KHz. I do not know how to test for HF instability. Would this be at rest or with an input signal?

The small signal response bandwidth is responsible for stability and for a properly designed audio amplifier it is 100s of kHz.  (1) The large signal response bandwidth is usually but not always just sufficient for audio because of slew rate limitations.  It actually takes some design finesse to get enough slew rate for power audio in a conventional design.

(1) If the designer naively implemented bandwidth control with the feedback network, then the measured small and large signal bandwidth is lower but this causes other performance problems so it is not favored.

[enut11]

I have completed most of the construction for calibrator #2.

Learnings from this build:
1) A bigger box was better but still not big enough. This one is a re-purposed modem box measuring 26x19x8cm.
2) The larger audio output transformer is a tight fit and probably too close to the font panel. There is an option of driving this externally from the LV output.
3) The small fan transformer was getting too hot so was relocated outside rear panel.
4) Some minor mods were made to the Hummingbird audio amp - 10uF input cap now 1uF and 1nF input cap removed.
5) 500 ohm input trim pot added to top of 220K amplifier input resistor

Next step is to test the signal output stability and noise.

[enut11]

Initial 1KHz tests of the new calibrator did not show any significant improvement over the the prototype wrt noise. See 10v and 100v tests below.

The new modules and transformers do, however, provide extended LV and HV ranges of 20v and 200v RMS.

I have yet to implement any form of feedback around the audio transformer as per previous suggestions (replies #42 and #44). I am not a circuit designer and would only be guessing what to do so I will leave that alone for the time being.

I am now going to concentrate on the input signal source. Previous tests have shown this to be the component that most affects the warm-up time.
I may also look at a PC based audio generator which is supposed to be very stable wrt amplitude over a wide frequency range (thanks @mzzj and @trobbins).
enut11

[enut11]

There are a few USB soundcards that have adequate audio bandwidth out to 80-90kHz before amplitude droop becomes noticeable.  The advantage of that type of oscillator source is that software can be used to null out harmonics from the 'output'.  As such, a very low distortion output can be sourced from the soundcard, but also extend further out to include any following buffer amp, as well as any step-up audio transformer.  Nulling harmonics in that manner can achieve low distortion levels that are intrinsic in just the ADC of the soundcard, and of course if the frequency is not too high in that bandwidth (eg. circa 20kHz fundamental for first few harmonics to be nulled), and the same applies for distortion introduced by a step-up transformer at the low-frequency end.  Certainly a cheap and easy way to start testing without the need for a low distortion oscillator.

Using a step-up audio transformer may benefit from some judicious loading of the high voltage output winding to maximise high frequency response flatness.  The output transformer likely needs to be 'hi-fi' in design to both push out amplitude variations from the first resonance, and to constrain the 'Q' of that resonance, given that resonance could well be below 50kHz depending on the transformer (a good transformer can push that resonance out to circa 100kHz).  The soundcard technique with suitable software (like REW) can even test the impedance of the transformer to confirm where the first resonance occurs (and hence the likely limit to frequency response for high voltage calibration efforts).

Hi @trobbins
I have an ASUS XONA7 U7 soundcard which should be suitable as an audio frequency generator. I have not used it in many years so I need to learn to drive it all over again. I originally bought it to test the distortion of amps and tape decks. You mentioned REW software as being suitable. Any how-to references would be appreciated.

[trobbins]

REW has a good help section.  REW has many functions and options, so it can take some effort to read all the relevant descriptions, and defer reading about features that you initially won't be interested in trying.  You would typically use the Generator and RTA functions for manual observation, and use the Measure function for automated sweeps.  The Preferences function is the starting point for selecting the soundcard and driver (hopefully there is an ASIO driver available) and the soundcard channels you want to use as output and input.   The https://www.avnirvana.com/ website has an REW forum, and is also where the software is available for download.  Some forums have a strong use of REW and threads such as https://www.diyaudio.com/community/forums/software-tools.123/.  Googling well known soundcards (like EMU0404, and Focusrite 2i2) brings up posts/threads where the soundcard has been tested using REW (and others) and indicates how they have been used and setup.  Googling tutorial may also be worthwhile - I don't look at the youtube posts but I can imagine there are a few helpful ones.

The 2019 review link (https://www.audiosciencereview.com/forum/index.php?threads/review-and-measurements-of-asus-xonar-u7-mkii-adc-dac-hp.8165/) indicates your card does have an ASIO driver available, but it sounds like there are issues, and the ASUS site only shows software up to 2016.  Hopefully you have a Mk2 as that has later software.  If you can get ASIO with operation at 192kHz at 24bit then that is great start.  Some soundcards are just fair performers, in which case you may need to spend time to characterise their performance for DAC and ADC paths to appreciate how to get a suitable performance for your application and to appreciate any limitations (eg. https://www.avnirvana.com/threads/measuring-frequency-response-and-distortion-of-amplifiers-%EF%BC%884x150w-rms-and-4x80w-rms%EF%BC%89.9764/).

If you are lucky enough to have a soundcard and ASIO driver that operates 'norminally' with REW, then that can open up a lot of measurement functions that imho far exceed legacy test equipment like scopes and distortion analysers and signal generators and impedance analysers/LCR meters, within the bandwidth constraints of typically 2Hz to 96kHz.

As an aside, I recently did a quick impedance check on a 1-to-9999 ohm 1930's 0.1% tolerance resistance decade box that I just restored, and due to the bifilar wind of the manganin coils and layering it seems like phase shift is fairly low out to 100kHz (circa 10deg at 30kHz).  So I hope to do some more investigation on how practical it may be to normalise for effectively zero phase shift up to 100kHz - all with the help of my soundcard and REW.

[enut11]

I installed REW on my Win10 PC and set the sinewave gen to 1KHz 1.0v out on the ASUS Xonar U7 soundcard and monitored the signal. Ambient temp was 28C.

The amplitude-time stability was poor compared to even my worst analog sig gen. It dropped by about 0.6mV per minute and showed no sign of leveling out even after 30 min.

Unless I am doing something drastically wrong, a PC soundcard source is not good enough for this project.

[Kleinstein]

Different sound- cards can be quite different in there performance. A super stable amplitude is normally not the main traget there, more like low distortion and stable frequency.

[David Hess]

Audio DACs can be very drifty as well.  Gain and offset precision is not a requirement for audio.

[1audio]

You would need to dive into the DAC implementation etc. to get a good sense of the causes of amplitude stability. I'll check a few later today (I have a number to check). Most audio DAC's derive their reference from the main supply. They may have an internal reference but its not super precise and may well be temperature sensitive and not compensated.  E.G. https://www.es.co.th/Schemetic/PDF/AK5394A.PDF  If you look the reference is sort of accessible but only for bypass caps. There may be an option for an external reference in the DAC on the Xonar card. Really good AC source stability may require a different approach (like the anal0g solution). My Boonton 1120 uses a state variable oscillator + sample and hold + Ref02 to get a stable source. The Fluke 510 also has a pretty carefully designed reference. It simpler and may be something that lends itself to the eBay oscillator board. https://xdevs.com/doc/Fluke/510A/510A_AA_imeng0000.pdf

[enut11]

I have completed the build of the AC Voltage Calibrator Ver2 and tested the LV at 10v and HV at 100v output. This final (?) version has a built-in 1KHz LD Audio Range Signal Generator mentioned in Reply #7 and has provision for an external signal generator.

Learnings from this project:
1) It is possible to build a stable 0-200vAC signal source at 100Hz to >30KHz frequency using readily available audio modules, voltage regulators and transformers.
2) Amplitude stability is almost entirely reliant on the signal source.
3) Low signal distortion is good but far more important is signal noise as this is amplified by up to x400.
4) Fan cooling of the audio amplifier is recommended and speeds up the settling time.
5) Separating the LV and HV output terminals allows the HV output to float above ground.

One application would be to calibrate up to 5.5 digit AC meters using a 6.5 digit or better reference meter.

[enut11]

As a follow up to this project, I found a cheaper solution for the power amplifier module.
A class AB MOSFET audio amp kit for less than $31AUD at:

https://www.ebay.com.au/itm/124926624992

This amp can be powered by up to +/-70v rails and deliver up to 40vRMS at the output.

As before, regulated rails improve amplitude stability. Also, for frequencies over 20KHz, the output stability capacitor should be reduced in value accordingly otherwise the 10R/1W series resistor will go up in smoke!

A heatsink is essential to assist with thermal tracking.

L7 mono MOSFET class AB audio amplifier board kit.
Specs:
Output power of 150W 8R DC +/- 56V rails (this applications only needs a couple of watts)
Frequency response 4Hz-350KHz, -3dB
THD <0.01% 100W 1KHz
Slew Rate = 38V/uS
SNR> 99dB
Requires AC 50V(or less)x2 transformer, or DC ±70V(or less) power supply.
PCB size = 78 * 63 MM