You should have a constant current source on that board to feed the zener some real, reliable current,
throw away that op177... ad706 is the way to go.
much lower input noise.
You will need to pay attention to a lot of details in the power supply in order to achieve the kind of accuracy you will get out of this reference, if you get it right. High spec op-amps and remote sensing will be needed to remove offsets and line regulation over load, respectively. You will also have to take into account the switching noise (mk. 3 is designed to be used with a tracking pre-regulator). If you don't filter the output right, your accuracy and stability will be down into the noise.
Good luck, I'm looking forward to this project :-+
Have you breadboarded this circuit? It looks like a comparator. How does it even start??
There is no clear starting current to get the zener into breakdown... perhaps the opamp starts hunting from rail to rail at the start and then it gets going? I am not sure. I don't like it this way, since you're not sure if it will start.
You picked an LM399, so you want stability, so let's give it some. You should have a constant current source on that board to feed the zener some real, reliable current, about 1.5mA or so. The LM399 can (at its worst) change its output voltage about 2mV for every 1 mA change in input current. The simplest constant current source is a constant voltage source and a resistor. So use your 15V regulated supply and a 5.6k resistor for about 1.45mA, and feed that into the top of the zener, it's not the best design, but it's ok. I say it's not the best because the 15V can change, according to the LM317 specs. i.e. it can change because of load, or temperature, or the 2 divider resistors that set the voltage can heat up and change, so your 15V changes too... When it changes, then your zener current will change (slightly), and the reference will change (slightly). How much it changes you will have to do the math :) If you are concerned about this, then the best love you can give your LM399 is to use another opamp to make a precision current source of 1.5mA and use that to feed the LM399. (there are many examples in almost every datasheet)
Once you have a constant current, then you can take the reference voltage from the top of the zener and buffer it with a low offset, low drift opamp. You can also use this same opamp to gain up your reference to 10V here, if you wish to.
The LM399 is very stable, but not accurate, so you'll need some way to trim the voltage to your desired reference. That's why I suggested the opamp to buffer and gain it up and at the same time you can trim it to exactly 10V.
The suggestion Free Electron made to use a AD706 is perfect, low offset, low drift, low bias, no chopper noise, and you get 2 in a single package, so one can be your current source :) But to make a current source you'll need another shunt reference, it doesn't need to be the same spec as the main reference, because when it changes there's a much, much smaller effect on the main reference.
You should have a constant current source on that board to feed the zener some real, reliable current,
His circuit provides a constant current source as stable as the reference.
I would be surprised if it doesn't start the OPA177 can't drive to ground. It probably sources some current while trying to, if not a bit of pull up on the op-amp output will fix that.
And yes if it is supposed to be a 10v reference it needs +/-5% of trim.
There are 4 purple colored rectangles at the pcb (1 vert & 3 horz), are they cut out holes ?
But to make a current source you'll need another shunt reference, it doesn't need to be the same spec as the main reference, because when it changes there's a much, much smaller effect on the main reference.
His circuit provides a constant current source as stable as the reference.
I would be surprised if it doesn't start the OPA177 can't drive to ground. It probably sources some current while trying to, if not a bit of pull up on the op-amp output will fix that.
I've not breadboarded it, but I'm pretty confident it will work. It is a constant current circuit. Initially, as the OPA177 output can't go to ground (typ. 2V), we will get an initial current of about 0.6 mA to the zener, through R4. It will start it up and get to about 6.9V. This voltage will be amplified by 1.468 to about 10.13V. Then the current will be (10.13 - 6.9) / 3k11 (R4) = 1.04 mA and will be as stable as the LM399.
Now reading the datasheets, why is the AD706 much better in noise than the OPA177? The typical input voltage noise @ 1kHz BW of the 706 is 15 nV/sqr(Hz) and the 177 is about 8 nV/sqr(Hz), both reading the noise graphs.
Given what you've told me about its startup, then I think I am also convinced it will get enough starting current and will eventually reach regulation. And I understand that the output buffer drops a voltage across the 3k11 and provides the drive current. But I see that design decision has problems that are easily remedied.
1) All of the regulating current for the reference comes from the output of the opamp, which is also feeding other circuits in your power supply. You should immediately see this is a problem for maintaining the stability of your reference. This would only be acceptable if you are going to put another voltage follower immediately at J2 before it goes anywhere else on to the rest of your circuits. And if so, why not put that buffer on the reference board itself, so that J2 is the output of a real buffer, not the output of your self-referential voltage reference that also provides it's own current source.
2) R4 provides a path for any voltage or current noise from outside your reference board to get in and throw the reference off course.
3) if J2 is feeding outside, and something outside sinks a large current, say an ADC sample and hold event, then it will steal current that is supposed to be for your constant current source to your reference. I don't know how you can call it a constant current source when you are taking that node off board and allowing other circuit nodes to siphon current away from it. That net feeding your LM399 should be stable and quiescent. Yes, I realize the opamp will provide more drive current when needed, but it cannot recover instantaneously, and do you really want the junction at the top of your current source changing that much due to dynamic loads off-board? I would not.
4) I don't like that your reference has no actually set current.. you really don't know what it will be, just trust that it will stop rising in voltage value as the opamp gives it more and more current until it settles.. and yes, I understand at that point it will be stable. But in my opinion I'd like to actually SET the current to something I specify, and maintain it there.
Actually, I'd guild the lily, and use the second opamp not for merely a current source for the main reference, but as a servo for the LM317 providing your 15V rail. servo the LM317 to stabilize it over time and temperature, and then you can set your 1.5mA current reference with a 5k42 low tempco resistor through the LM399. Use the original opamp as a buffer amplifier where you can dial it up to 10.0000V
One of the great things about Dave's forum here is that there are great discussions that get one thinking..
I was curious about this discussion, the AD706 vs the OPA177.. So I did some math. I've attached images and Smath Studio output for you viewing pleasure.
I don't follow you there. The current is set by R4. Why are you worried about the turn-on transient current if it every circuit will have the same problem? Even the best constant current circuit will do exactly that as it turns on. In the end, it won't matter, as I intend to keep it on as long as I can.
The circuit, as you drawn is pretty much the same as my circuit, using the LM317 as a high output current buffer. If you put another buffer in there, you will loose that.
Where did you get those bias drift numbers for the OPA177? The datasheet, page 2, (http://www.ti.com/lit/ds/sbos008/sbos008.pdf) lists 85 pA/C Ios drift and 60 pA/C Ib drift, a lot worse than AD706 but much better that the ones you've used.
throw away the socket... you can get electrical noise due to stress in the socket....This is a typical "once heard" statement used as an answer for all questions. Stress is less a problem also because of the CerDIP package, air flow is much more critical. But you seem to forget the LT1001 is not the reference but the buffer amp, so I don't care about your "electrical noise due to stress in the socket" argument.
Zeners are current driven objects. The input cuurent noise is in the femto ampere range for an ad706 and i. The picoamp range fro the op177. Furthermore the ad706 is dc stable. This opamp is built for dc operation. The input bias current is also stable over temperature, that of an op177 is not.
This drift in n put bias current willl shift the operating point of the zener ever so slightly.
BTW: Resistors can be bought over here http://www.rhopointcomponents.com/components/resistors/precision-through-hole.html (http://www.rhopointcomponents.com/components/resistors/precision-through-hole.html)
Thank you Branadic, I got this book and it already payed for itself...Pretty good stuff isn't it? I still read serveral chapters from time to time.
You should consider to stabilize the heater voltage of the LM399 instead of using a (unstabilized?) 18V supply. I have observed 0.5 - 1.5ppm/V output voltage change between 10-16V heater voltage.
As I said in the other post, I don't really care of the accuracy. I will trim it elsewhere in the power supply. I just want it stable.
The two circuits discussed in this tread so far seem to be the self bias version and the staged version. What do you think about the "combined voltage/current regulator" version vs. the "self bias" version (compared to the simpler staged version)?Hello,
But the resistor arrangement around the op amp on the output as either a voltage divider + voltage follower or a non inverting DC amplifier does not change the operating modus of the circuit.The resistor arrangement has the most influence on the stability of the output voltage. I recommend highest quality precision resistors (either precision wire wound or Vishay metal foil) for this. And yes: they will cost more than the reference itself.
I was more interested in having opinions on the principles for the 3 solutions.
I did not know that the heater was not "purely resistive".
I think a very good design consist of a cascaded LM317 arrangement feeding the heater and a temperature compensated constant current source for the zener (e.g. LM334). Together with a chopper amp such as the named LTC2057 and some pi filter to limit the whiteband noise the design could be completed. I would agree that its worth setting the ouput voltage to 10V nominal and divde them with LTC1043 to 5V or other values.
What do you mean with cascaded arrangement? (schematics).
QuoteWhat do you mean with cascaded arrangement? (schematics).
"LM317+cascaded"
http://www.acoustica.org.uk/t/3pin_reg_notes4.html (http://www.acoustica.org.uk/t/3pin_reg_notes4.html)
Of course a good designed ADC or meter will add additional mains line frequency reduction by choosing appropriate integration time.
This is why old meters would have a jumper/front panel setting, or would even have different crystals, for 50 Hz and 60 Hz. Modern meters tend to use a PLL from the mains frequency to produce the sampling clock.
Why not choose a multiple of 100 ms as integration time?That would work. 10 samples per second is a fairly slow sampling rate, though, compared to the 50/60 S/s that are possible with 1 PLC.
This would give multiple of 5 PLCs on 50 Hz and multiple of 6 PLCs on 60 Hz.
Ok the LTC2400 has to be configured by PIN to 50/60 Hz suppresion (> 110 dB) at Pin 8.This pin indeed controls the internal conversion clock to be a multiple of 16.67ms/20ms.
When connecting to VCC the integration time seems to be 80 ms.
(can be measured as spikes on the input of the LTC2400)
More later...
and interesting parts hidden in pink foam)
Has anyone tested if the LM399 also has some kind of zero tempco current
(near the heater temperature) in the valid range of 0.5 .. 10mA for the zener?
I really dislike buying schmartboards and soldering TSSOP/SOIC packages...
Noise measurement of triple LM399H with constant current supply and simple filtering (see pictures above):
Pass No1: 0.88 uV p-p;
Pass No2: 0.87 uV p-p.
Do i need a separate reference output for the DAC (REF which is a bufferd output of the LM3999) or could i use the 10V output. I thought i build a Voltage divider(switched Capacitor with an LTC1043) to feed the DAC.
The 9k and the 20K resistors need to be low TC ones. The 100 ohms and the 1k Resistor aren't so critical is this assumption right?
Is it better to use one 14 or 16 bit DAC or two (cheaper) 8 bit DACs.
I had about 50 of these LM299 in my junk box and figure with not do something with them.
This is interesting to know, maybe you want to tell more about what you are planing to do (circuit desciption)?
Why LM399 and not LM199AH/883 with higher temperature specs?
- LTC1043 as divider (*3/4 = 5.2 V reference)Is that one or two 1043s? If you got 3/4 with just one IC, I have a puzzle to solve :) If two, how did you opt to join the *3 and /4 stages?
Is that one or two 1043s? If you got 3/4 with just one IC, I have a puzzle to solve :) If two, how did you opt to join the *3 and /4 stages?
Is that one or two 1043s? If you got 3/4 with just one IC, I have a puzzle to solve :) If two, how did you opt to join the *3 and /4 stages?
I tried to get 10/7 working in LTSpice, but I couldn't get the sim to be stable.
Damned: I should have patended my single LTC1043 *3/4 solution
The circuit uses a 200k resistor in series with the reference zener diode. Does anyone know why they used 200k which would set the zener current between 25 uA and 55 uA.That 200k isn't setting the zener current; the opamp output is driving the zener through a 5k resistor.
Let me take the role of the "Bild" Newspaper:
SLOTS IN PCB INCREASE AGING DRIFT
Effect discovered by amateur scientist contradicts prior scientific work considering slots to be advantageous. We all gonna die!
So, it seems that the slots are counter productive in this use case. Maybe, it would be better to let your LM399(s) be on a separate PCB from the rest of the circuitry. That would make the whole LM399 assembly easier to insulate.
Hello macfly,
not so good as in the LTZ1000.
you can see this in my tilting experiment.
https://www.eevblog.com/forum/projects/lm399-based-10-v-reference/msg360779/#msg360779 (https://www.eevblog.com/forum/projects/lm399-based-10-v-reference/msg360779/#msg360779)
Further the stability is dependant on heater supply voltage.
You can get good results with stabilized heater voltage and good thermal management.
With best regards
Andreas
Now you understand my first question about the stability of the regulation?
Now you understand my first question about the stability of the regulation?
Hello macfly
Not really. (how are the exact measurement conditions and what do you mean when speaking of temperature coefficient).
The unheated reference should be in the around 50ppm/K range. (Similar to the LTZ1000).
The heated reference near room temperature is much better.
That brings me to the idea to select the references according to raw tempco.
It seems that it will be an even better reason for me to follow this thread, my 6.5 digit meter had a 200uv offset at 10 v.
What are the differences in the H vs. AH specced parts?The A-type is selected for a smaller (overall) tempco and smaller initial tolerance (National Semiconductor only). See datasheet.
Hello macfly,
the results are too good to believe.
That brings me to the idea to select the references according to raw tempco.
I have looked up my measurements: unfortunately I did not document the values carefully.
But the difference of 2 of my references is at least 4-5mV unheated / heated.
@macfly + Mickle:
Did you use National Semiconductor parts or LT parts?
On my side all are National Semiconductor parts up to now.
@macfly:
Did you kelvin sense of the zener or is the heater current affecting the zener voltage?
With best regards
Andreas
Interesting that the lower TC modules tend to be < 6.9V. Makes me wonder if the others would have a lower temp Co at a lower current.
in my opinion the shelf life stability is the more interesting spec.
(rarely found in the data sheet).
Usually you would not use a precision resistor near the rated power.
I have done the correction calculation for a linear ageing (linear regression coefficient).
The correction is applied on the averaged measurement values and not for the original 1 minute values.
The spread values with the foam do not really change:
CH6 foam corrected: 4.4 uV = 1.3 ppm (unchanged)
CH7 foam corrected: 10.4 uV = 3.0 ppm (was 10.1 uV)
The spread values for the "cotton" isolation seems even to be better for the slotted PCB:
CH6 cotton corrected: 4.6uV = 1.4 ppm (was 4.1 uV)
CH7 cotton corrected: 2.4uV = 0.7 ppm (was 3.6 uV)
remember that standard deviation for my setup is around 0.25 ppm or 1 uV.
@turbo
R43 and R44?
but I think the more important function is to act as a heater itself to heat the copper trace.
Interestingly, you'll see the stand-offs for the big shunt uses intentionally thinned out pattern to connect to ground. I think this is to minimize thermal transfer from the trace.
But the output voltage of LM399 has a relative large dependancy on heater voltage/current.
I've heard this before.
I've heard this before.
Perhaps in one of my threads.
Paraffin is only for temperature equalization. S5-61 foil resistors have a significant TC difference.
I couldn't get the 7.5K for the heater , but have some 6.49K 5-ppm resistors.
Can i use one of those for the heater ?
In this case a 6.5K will do its job.
You will have slightly more than 1mA at the zener.
-> ageing drift increases slightly.
Andreas
my EE Guru said it was a good idea.
my EE Guru said it was a good idea.
The standard cirquit would tap R3 on the other side of R5.
So voltage loss due to R5 is canceled out.
Usually R5 can be choosen around 100R or even somewhat below
without loosing stability with large capacitive loads. (C2 necessary).
With best regards
Andreas
If you remove all the small green junctions where a single wire simply connects to a component, it would make me very happy :)
Absolutely keep them for three-way junctions. If you are drawing this using "Wire" in Eagle... don't! Use "Net" instead, and the junctions will for the most part automatically show up when they are needed.
Side note: 0.1 ppm/C is only after selection. All resistors in my reference have a 5 ppm/C class (max tempco). Typical is 10 times better. By selection is possible to achieve 0.1 ppm/C (20-50 C range).
Hello Mickle,
would you let the cat out of the bag, where we can buy S5-61 foil resistors?
My version of the quick-and-easy LM399 voltage reference :)
Mickle ... Easy
Is that transistor arrangement for controlling the Zener from the 10v , meaning super stable ?
Why the transistors , to avoid loading of the Vref Out ?
From where can I get those tiny dewar cylinders.Hello Andreas,
Do you have a source with reasonable pricing?
Blue (K2015) is the LM299 , strange that the LM299 (Board and/or trimpot) seems to have a negative tempco.
Blue (K2015) is the LM299 , strange that the LM299 (Board and/or trimpot) seems to have a negative tempco.
Interesting would be also the raw value of the voltage.
Which type of trimmer do you use?
AccuTrim series of vishay?
Otherwise you should use the "standard precision" trimming scheme shown e.g. in the AD587 datasheet (figure 3+4).
extra 10-100k trimmer from the output to ground and the tap via a "precision resistor" (RT in datasheet) to the center tap of the 9+20K divider.
with best regards
Andreas
raw valueIs that the LM299 Vout ?
Not perfect , but maybe usable
Would be interesting if a thermal coffee mug
(stainless steel with vacuum) gives similar results.
Hello Mickle,
would you let the cat out of the bag, where we can buy S5-61 foil resistors?
Hello,
You could also order some VHP100/VHP103 or from the HZ series resistors from vishay and select for TC.
Most probably the S5-61 resistors are no longer manufactured. Otherwise you could google after them.
with best regards
Andreas
If I'm remembering correctly, the LM299 has a negative tempco.
When I look for dewar I only get to pages with LN2.
And the lowest cost dewar is around one HP34401A.
$10 sounds reasonable.
The replacement glass part might save a few € against the full part and also the life of a poor nice oscillator !
(see 9.032 011, 100ml, also available without the blue jacket, so the price could be somewhat lower)
I found dewar available from different suppliers:
The replacement glass part might save a few € against the full part and also the life of a poor nice oscillator !
Rather, I'll achieve trim by adjusting the current driven through the LM399 zener.Good Idea: so you will have more fun at adjusting: Trim a bit, wait 15 minutes for thermally equalizing. Trim again. w. w. w.
Using this setup, a drift in the LM317 of +/- 0.1V results in a current source error of only 2.3 uA. Not bad!Too much: with 1 Ohms for the zener this is 2.3uV output voltage change.
So I tried wrapping my brain around making a high-side current source, and this is what I came up with:The LT1001 is no rail/rail OP. The cirquit violates at minimum the input common mode range below 2mA.
Requirement of datasheet is 1mA +/- 0.1% for long term stability.
Read the first post of this thread. It's a better way to create a constant current as stable as the LM399 it is powering.
Read the first post of this thread. It's a better way to create a constant current as stable as the LM399 it is powering.
Does anyone have experience with using current regulating "diodes" for this purpose?These are simple J-FETs where Gate + Source is connected together. -> you will never satisfy the requirements of a precision circuit.
There is a large possibility that the tempco is not linear but either a parabolic or s-shaped curve (due to compensation of the zener with a Vbe).(https://www.eevblog.com/forum/projects/lm399-based-10-v-reference/?action=dlattach;attach=99065;image)
Has anyone tested if the LM399 also has some kind of zero tempco currentI have tested 0.6, 1.0, 1.1, and 1.5mA. The resulting graphs were effectively identical.
(near the heater temperature) in the valid range of 0.5 .. 10mA for the zener?
My range of unheated tempco appears to match Andreas exactly. Were those units also LM399A?
I have tested 0.6, 1.0, 1.1, and 1.5mA. The resulting graphs were effectively identical.
1N829A Zeners have their minimum T.C. around 7.5 mA +/- 3 mA
p.s.: do you have a photo from your measurement?
I've got some data on the LMx99/A series from Bob Pease, as soon as I can dig it out of my archives I will try to post it, may be difficult in the form it is.Neat! I look forward to it. I did get the impression he played around with this part a lot.
I just can´t imagine how your "ageing oven" from night lamps and tiles looks like.My pleasure to do a bit of show and tell :)
sounds interesting.
Certainly an interesting build, how stable can you get it?It's not regulated at all; it varies over a 5C range as ambient conditions change. Good for aging parts, not so ideal for accurate TC characterizing.
good enough, i might try it in the future.Certainly an interesting build, how stable can you get it?It's not regulated at all; it varies over a 5C range as ambient conditions change. Good for aging parts, not so ideal for accurate TC characterizing.
I may update it some day with a peltier module+fan instead of the lamps and make controlled environment out of it. (But peltier modules typically don't like going to 125C, hehe)
My pleasure to do a bit of show and tell :)
The maximum TCV of a LM399H is 2 PPM/°C, if you are seeing anything higher than this, there is a problem with the test setup. The LMx99 is very linear, regardless of operating current, the internal temperature is regulated at ~90°C. The operating ambient temperature range for the LM399/A is 0°C to +70°C. The internal temperature of the LMx99/A is self regulating within the specified ambient temperature range. Your measurements should indicate a linear TCV with change in ambient temperature. All those curves seem to indicate problems with the test setup, even with repeatability of the curves, it indicates the error sources are relatively stable themselves. Yes, the LMx99/A run at 90°C so you are going to see heat radiating off of them, the zener generates much less self-heating as is the design intention. The heater runs at about 300mW at 30V, the zener would run at ~70mW at 10mA, at 1mA only ~7mW. The heater compensates for the zener's heat as well, mostly leaving just the TCV of the zener itself. The drift spec indicates a zener current of 1mA +/- 10%. I have used the LMx99/As and I have never seen anything beyond a consistent TCV with them. With varying ambient temperature, I have measured nothing more than the drift of the zener, usually under 1 PPM/°C for a LM399/A version. Thermal EMFs can be really difficult to control and they tend to be nonlinear in nature.
Hello jpb,Again, as I'm not Galaxyrise I can't claim any deep knowledge, but my understanding was that by finding which devices had flattest curves around 90C good devices could be selected without having to monitor the heated devices for a very long period - but this is just a guess on my part. I think the point was to find devices (and current levels) which would be optimal when the heaters were used in the final circuit.
The point of using an LMx99/A is to utilize the internal heater to stabilize the reference zener's operating temperature, hence the exceptionally low TCV. Attempting to measure the LMx99/A's zener TCV without the heater energized is an invalid test of the part and only erroneous readings will result if that is the case.
Attempting to measure the LMx99/A's zener TCV without the heater energized is an invalid test of the part and only erroneous readings will result if that is the case.
I have attached Bob's note to me
plus a National data sheet from 1999 which gives some interesting specs. I also attached the LM199 data sheet from Linear Tech from 1990. Linear made some improvement in the TCV over the older National part, but we're talking a few tenths of a PPM/°C here.The NS sheet does have interesting extras, like that 1000hr aging chart. And a TO-92 part, teehee!
Your assumption about the 'zener' diode in the LM399 is understandable but it is still an invalid assumption. ... Outside of that temperature range, the buried zener will have a composite TCV curve made up of the entire voltage reference circuit not just the buried zener.Characterizing the entire reference circuit is exactly what I thought I was doing (as it's what matters in the end), but I hadn't really thought about that in detail. I was just reading through National Semi's App Note 161 (http://www.ti.com/lit/an/snoa589c/snoa589c.pdf), and I can really see what you mean. The current changes, especially, basically don't affect the zener at all! (Though it does affect the circuit as a whole.) I suspect I will run a similar experiment when aging is done, but with the heaters on (and a narrower temperature range, obviously.) It will be interesting to see if it correlates to the unstabilized data at all, or if the charts I produced are just a curiosity.
If you have several LM399s available, you could parallel them using the appropriate summing circuit to get an improved composite characteristic, it doesn't just work for reducing noise.That's the plan! I've seen where Pease talked about running them in groups of 4, and I was figuring 4-6 based on how they looked in the latter half of aging.
I have found one note that Bob Pease sent me so far, you may have a hard time reading his hen scratch, this was part of a letter he sent me back in 2003, he did not mention just what meters were involved in the test nor the exact date of the test. The 'graph's (I'll use that term loosely) indicated a repeatable drift in the measurements of the LM399 (he did not mention what the suffix was, an H or AH). The test was over some months long but Bob didn't mention just how long, usually a long term drift measurement was about 10-12 months.
Note to Ed Pettis
If you look at the LM399’s you might see a PATTERN for the first couple weeks, and the last week:
(figure)
- Now if they all look like this - is that plausible? (No)
- Is it likely that the measurement system’s Reference V drifted (figure) ? Yes.
We keep alternative references and if we see that everything is drifting like (figure) then we know these parts are really quite good. If we had to, we could crank in some corrections and show that these parts are actually more stable than the measurement system! But as we are only talking about ~ + and - 7 PPM.
Hello,
Characterizing the entire reference circuit is exactly what I thought I was doing (as it's what matters in the end), but I hadn't really thought about that in detail.
or if the charts I produced are just a curiosity.
I think the shift of the top of the TCV curve in the last graph is due to self-heating of the LM399. If you turn on the current and take a voltage reading then turn off the current, and allow some thermal settling time between measurements-- I think these shifts will go away.I started the 7.5mA and 9.5mA tests by letting the references heat the oven to get a sense for the self-heating, but now that you suggest it, I'm pretty sure I didn't compensate for it nearly enough. Your measuring method sounds like a much better way to eliminate the error. It would be harder to do over the entire range, but so long as the parabola is still true, I would only need to spot check a few points to get a rough sense for the shift. I will endeavor to do that this weekend.
Since the internal circuitry of the LM399(A) shunts extra current around the Zener, there is no way to decrease the awful noise spec by increasing current-- so you may as well run it at the 1.0mA current to get the best long term stability.
I used 3 each tantalum capacitors in parallel, they were 68uF/16V, they couple the input from the device under test to a 25 ohm 0.1% 25 ppm resistor to the inverting input.Cutoff frequency is 31.2 Hz, but not 0.1 Hz.
Hello,QuoteI used 3 each tantalum capacitors in parallel, they were 68uF/16V, they couple the input from the device under test to a 25 ohm 0.1% 25 ppm resistor to the inverting input.Cutoff frequency is 31.2 Hz, but not 0.1 Hz.
Attached is a picture off my board.
I would remove the socket of the first input stage to further reduce the noise.
And its always a good idea to do some thermal isolation (e.g. a piece of cotton or cloth)
I think the shift of the top of the TCV curve in the last graph is due to self-heating of the LM399. If you turn on the current and take a voltage reading then turn off the current, and allow some thermal settling time between measurements-- I think these shifts will go away.Confirmed. I took spot measurements to locate the peak of the parabola, and it appears to be right around where it was at 1mA. I'm going to remove the errant graphs lest they confuse someone who doesn't read the whole thread
although the discussion gets now rather off topic:
With best regards
Andreas
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It´s your decision.although the discussion gets now rather off topic:
I have removed my postings from this thread.
After my conversation with Bob Dobkin, my take-away was that there is nothing wrong with building a voltage reference around some paralleled LM399A's [or even just one of them really]. The more you parallel, the lower the DC-10Hz noise will be. Because they run so hot, hysteresis is not much of an issue, and after a long burn-in period they are only going to drift 1-2ppm/year [if left on 24/7], which is really not that bad. Additionally, there is an opportunity to build a voltage reference that has no batteries [or at least the batteries don't need to hold the reference circuit up for more than a few hours]-- Bob said that these devices don't drift if they are off. So, you could build a voltage reference that you leave off most of the time, and then only turn it on an hour or so before you are going to use it [or calibrate it]. In this use case, the voltage reference could hold it's calibration within 1ppm of the SI volt for many years, which is important for hobbyists, because calibrations can be rather expensive [especially if they are using a JJA].A good answer to the starting issue is to use a fet with a significant gate threshold, thus the circuit is already starting to receive zener current even when the output is pegged to low. this also reduces the current demand and the heat issue in the opamp that is required to have low drift and shifts in output.
My burn-in recommendation for the LM399(A) is fairly radical-- it would involve building a very well insulated oven that was controlled at 150C [+/- 5C]. The LM399(A)'s are placed in this oven for ONE YEAR [and now you can see the need for excellent insulation!] This will provide "artificial aging" that approximates about 67 years of natural aging, and could result in references that drift less than 1ppm/year, even if you left them on all the time-- and much less if you only turn them on when you need them. To "soften" the die-attach [which can cause sudden jumps of 1ppm or so-- like a "pop"], the references are placed in a live circuit [no need for accurate resistors in this circuit], and then that is placed in a freezer, and the power is cycled on/off [one minute on 1 minute off] for 90-days. This way, if there are any bubbles or micro-cracks in the die bonding material, these will be "worked out" over this period. For obvious reasons, it is more economical to use this burn-in procedure on a hundred or more references at a time-- and because it takes so long, one has to have great patience.
Bob Dobkin said that you should parallel at least 6 of the LM399(A)'s, but I think 4 of them would be sufficient, and this is still less cost than an LTZ1000(A) based reference. You get a sqrt(N) reduction of noise, so this would economically reduce the noise by a factor of 2. The next level would be 9 devices for a noise reduction factor of 3, and then 16 of them for a noise reduction factor of 4. 9 devices would exceed the cost of an LTZ1000(A) reference, so 4 LM399(A)'s is about the economic limit for this technique.
Bob also said that the current limiting resistor for the LM399(A)'s Zener should be tied to the stable 10V output. This causes a start-up problem, but in the two threads I think this issue has been solved in different ways-- and all of them should work.
In the LTZ1000 thread, you can see one of my early designs of a PWM circuit for the 7V-to-10V boost circuit. Since then, I have refined the technique and it is very much simplified [using only one 32-bit PWM stage, and one filter]. No critical resistor ratios or absolute resistor values are needed, and thus there will be no resistor-related drift [at least in the boost circuit]. I have some more work to do on the digital side of things, and of course I need to build and test a statistically large enough population of these references-- but once I have done that successfully I can post my findings. Since the boost circuit could apply to either an LM399(A) based or an LTZ1000(A) based reference, I will probably start a thread just for that and link to the two voltage reference threads.
Hello,
Just another updated ageing chart of the CH6 + CH7 LM399 devices.
If you remember this was the "slot or not" PCB of branadic.
The drift is now scaled in ppm.
...
With best regards
Andreas
I went through the whole thread again, but could not find any description, how your monitoring system works, and on which volt reference it is based.
Therefore, I could not draw any conclusion, which drift you really display here, i.e. really the drift of the DUT, or the drift of the monitoring system.
Would you mind explaining your practical setup, and how you are able to determine the absolute drift of the LM399s?
Thanks
Frank
Hello Frank, Ken,
...
ADC#13 is a well aged LTC2400 based device with a AD586LQ voltage reference and a temperature sensor.
T.C. is compensated by a 3rd order correction curve.
..
ADC#13 stability can be found by the following charts of my "daily" automated measurements.
These attached charts are done with offset compensation.
Offset is measured once at beginning of the measurement and subtracted from the following readings.
If I set day 0 at the beginning of LM399 CH6 + CH7 measurements you can see that during same time
ADC13 does not drift more than about +/- 1 ppm against two LTZ1000A references (blue and green)
which is also negligible against the 10ppm of the LM399 CH7 reference.
For the LTZ1000A references I try to get reliable calibrations.
But the only thing that I can say from comparisons to other instruments like Keithley 2000 with calibration history
or Fluke 5520A is that the drift of the LTZ devices is up to 2 ppm/year against those devices.
But I cannot tell wether the calibrator drifts or the LTZ1000A.
I will still need some years to make a final decision.
With best regards
Andreas
Hello Andreas,
although the AD586 is quite mediocre concerning TC and ageing, your setup with TC compensation and comparison (frequent calibration?) against your LTZ1000As gives good stability / uncertainty for your ADC#13, I think.
.....
What do you think about that?
Frank
I am also interested how is switching done between different channels?
What is needed, is a traveling LTZ standard, and a group of volt-nuts willing to do the comparison.
What do you think about that?
Frank
I am just starting to read this "LM399 based 10V Reference" Subject,,,, am on pg 7,, But am wondering what spice model anyone might be using for the zener.. Although I use Proteus,,, I don't see anything in LTSpice on the LM399.... You all have done such a good job describing the device's in's and outs... Never seen so much activity for a 4 terminal device... Any response on this will be most appreciated..
Wonderful then,,,,, Any thoughts on using a the "Portable Calibrator" circuit WITH " Walt Jung's, Analog Devices, Build An Ultra-Low-Noise Voltage Reference... Electronic Design 6/24/93".... Of course substituting both op-amps with the LTC2057... Like everyone in the world, I'm looking for low drift,, low noise... all in a 3 terminal package...aaahhhhhhhh. Any thoughts will be appreciated, thanks...
I did not understand how the flicker or 1/f noise of the reference is measured. I could understand that a chopper amp could reduce the impact of the needed voltage amplification. But the amp could only amplify a voltage difference of two potentials or one potential to a ground reference point. But the reference is the DUT itself have an offset of 7V against the chopper amplifier. So there should be second reference which significant lower or well characterized flicker noise to counter the DUT reference voltage. The difference should also be smaller than the requested gain of the measurement.
So what kind of offset source is used?
I did not understand how the flicker or 1/f noise of the reference is measured. I could understand that a chopper amp could reduce the impact of the needed voltage amplification. But the amp could only amplify a voltage difference of two potentials or one potential to a ground reference point. But the reference is the DUT itself have an offset of 7V against the chopper amplifier. So there should be second reference which significant lower or well characterized flicker noise to counter the DUT reference voltage. The difference should also be smaller than the requested gain of the measurement.
So what kind of offset source is used?
I use a large (3200 uF) electrolytic capacitor selected for low leakage current.
Noise floor is <0.2 uV together with a 1K input impedance and a LT1037 OP-Amp
in the first amplifier stage.
See also AN124 of Linear Technology.
With best regards
Andreas
I have further doubts that the dielectric absorption set the lower limit above the reference.
From the pairs you can calculate the indidividual noise spectra.
I doubt that one can calculate individual noise from pairs.
Indeed you can :) Click :
www.intersil.com/data/an/an177.pdf (http://www.intersil.com/data/an/an177.pdf)
What you measure with the pairs is the sum of the noise powers
What you measure with the pairs is the sum of the noise powers
Hello,
what do you want to measure?
wideband noise > 10 Hz or
1/f noise 0.1 .. 10 Hz or
drift (below 0.1 Hz).
For wideband noise you are right: the power (effective value) adds.
The voltage adds only geometrically. (square root of the squared sum).
For 1/f noise you usually can only measure the peak-peak voltage value.
And if you really have done such measurements you now that
from measurement to measurement you have a lot of variation which
makes it practically not possible to measure differences.
Except when one of the 2 references has much lower noise.
-> my suggestion to average at least 9 references as
"low noise reference" to have less than 10% error for the DUT.
With best regards
Andreas
LM399H with the external oven :D
- TempCo ~ 0.25 ppm/C;
Which ADCADC - Mark-space type with convergence accelerating signal.
which Reference
which processor.
My mother today emailed me that she considers the LM399 thermography movie by Andreas and branadic "art" :)
FYI there are LM399 (NS) on the bay at $4.60. I bought some a few months ago and they where OK, even if I have not done any deeeeep measurements on them. Seller Polida2008. I have no affiliation whatsoever.
A temperature difference of 1°C between the two leads of the reference will generate about 30 ?A.
Keep lead lengths short.
Ok, I have finally evaluated the measurements = .csv tables of Branadic from the LM399 thermograpic measurement.
Long leads without slot: around 5 degres stray within pad + 1,7 degrees from pad to pad
Long leads with slots: around 5-6 degres stray within pad + 2,3 degrees from pad to pad
Short leads without slot: around 9-11 degres stray within pad + 3,1 degrees from pad to pad
short leads with slots: around 12-15 degres stray within pad + 6,6 degrees from pad to pad
So from the measurement the conclusion is that with long leads the stray (=thermoelectric voltage) within pad is lower than with short legs. And the thermoelectric voltage between different pads is lower without slot.
If there should be mechanical issues the slots should be at least around 15 mm from the reference to give the pads the possibility to equalize the temperature.
FYI there are LM399 (NS) on the bay at $4.60. I bought some a few months ago and they where OK, even if I have not done any deeeeep measurements on them. Seller Polida2008.
To your sudden die of the reference:For the charts I produced awhile back in this thread, I had the 10 LM399 in series driven by a 1mA current source, so they'd all have the same current during measurements. I had my DMM switching between them via switching card. I ran like this for days.
It is winter. Perhaps you have zapped it by ESD.
(I have always to think of the "sweet spot voltage" between unheated and heated device.)The resulting average is 7.0689. None of the units I got were under 7V. I didn't record normal operating voltage on them individually, but I believe they ranged from 7.05V to 7.08V.
I tried to improve the resolution of this measurement by using another 7V reference on a different supply.
10 Zeners in series (70V) would also violate the max 40V spec of the internal diode if the heater is supplied from a common source.They weren't. I did try to take that 40V spec into account, at least, and the failures were closely correlated to the DMM scanning and not the static setup (which had been running for days before I did the scan.)
What kind of supply?Both the LM399s and the 7V were on linear supplies.
And also linear supplies may have some 10 to 100 pF (parasytic) capacities.Some capacitance-related answer certainly seems likely. Parasitic to earth ground, you mean? Hmm... Perhaps I'll try to recreate it with resistors instead of LM399s and see if I can track it down. Seems likely to be educational at the least! I don't want to clutter up this thread with it, though; I'll put something in Beginners if I do play with it some more. Thanks for your input!
Zener current is set to 4.5 mA ( in datasheet it is set about 1mA )
Wow, cool. You did more progress on this module, that I had ;DThanks, I do not think so, I also needs to spend more time on LTZ1000. For measurement I needs to finish the climate chamber and some MUX. I would like to assemble additional three PCB with LM399AHs.All with Vishay UXB resistors and same enclosure.
Would be happy to see it in good use with some measurements.
Zener current is set to 4.5 mA ( in datasheet it is set about 1mA )
There is no advantage in setting the zener current above 1 mA.
(see statement of Ken).
Maximum stability of LM399 will be at 1 mA.
With best regards
Andreas
@Plesa #334,
You should be using either Bourns 3250 or 3290 wire wound pots (these are military qualified) if you want long term stability, film type pots are not nearly as stable over the long term (2-3 years compared to decades) and also require a minimum amount of current to flow through the wiper to keep it in 'good shape'. The WW pots do not really have to have any current flowing in the wiper but 1uA-3uA is suggested. The Vishay 1280G pots may have a bit better resolution but they are not nearly as stable long term and also cost a lot more than the WW pots.
The main drawback of the LM399/A references is noise compared to the LTZ references, about 4-5 times higher but at the same time they can be just as stable over time as the LTZs. If your application does not need the lower noise level of the LTZ then the LM399/A is certainly the lower cost way to go. As Bob Dobkin said, the higher current level does reduce noise but increases drift, there is no way around that tradeoff, if a little higher drift is okay then you are set with the higher current but noise is a very important spec and sets the minimum accuracy floor of the reference, you cannot filter the noise out from a zener so if you need low noise you must start with a low noise source.
I surprised how stable it is during burnin period (<1ppm).
Price is much too high. (Compared to new LT parts from DigiKey).
what do you think about this offer:
Your opinions?
So it may be better to buy this, slightly more expensive but pre-aged ones:
Wauu, interesting piece of equipment. MickleT are you going to share schematic?
DC-Cal Lite v1.0 - 10V/10mA/1mA calibrator, based on the AD5791B DAC.
0.1-10 Hz noise: 0V - 0.4 uV p-p, 10V - 4...5 uV p-p.
0-10 V INL <1 ppm (measured with Solatron 7081 DMMs).
Output resistance 0.003 Ohm (DC voltage mode).
TempCo ~ 0.1 ppm/C.
taking data each month to calculate the drift for each of it.
So how exactly did they select the voltage references:
Including the measurement of the noise?.
(current limiter maybe ?)
if this lowering temperature also work with LM399 could somebody share their experience and how to do it (current limiter maybe ?)
(current limiter maybe ?)
Hello,
this is no good idea and will lead to instabilities.
(how do you get the inner temperature of the LM399?)
The only practical way is to leave the heater unconnected and put the LM399 in a own oven.
You could also use the LM329 (= LM399 without heater) which is cheaper but unfortunately no longer sold in metal can case.
With best regards
Andreas
I suspect that the noise level on an unheated LM329 should be considerably lower than on the LM399 at 90C.
I suspect that the noise level on an unheated LM329 should be considerably lower than on the LM399 at 90C.
I suspect that they have only "beautified" the diagram.
25 deg -> ~ 300K
85 deg -> ~ 360K
why is sqrt(300K) = 17.3 significant lower than sqrt(360K) = 19
when aproximating the noise as Johnson noise?
with best regards
Andreas
You could also use the LM329 (= LM399 without heater) which is cheaper but unfortunately no longer sold in metal can case.
With best regards
Andreas
Johnson noise calculations won't work here. Just have a look at the noise density graph available in the LM399 datasheet for 25C and 90C - the increase is about 1.5 times (and most likely more in the low 1/f region).
Cheers
Alex
So you should be able to tell which are with heater active and which with heater disabled.
$15 for a half LM399 is a bit too pricey for me.
With best regards
Andreas
1) No, I can not tell the difference, sorry! Which means the datasheet is not to be trusted ;D .
Also LN2 temp is not that stable, it vary about 0.1K over minutes. Dewar is not pressured, so likely LN2 getting liquid air and oxygen causing temperature to vary.
I'm more concerned about voltage zener reading, not the temperature itself.
Have Honeywell HEL-705 1k RTDs so can use those.
So I would definitely prefer something like 40-50 C so well above ambient, but not as high as the oroginal LM399.
Hello branadic,
LTZ#2 was intended to show the drift of the ADC reference.
LTZ#2 itself has most probably a drift of around -2 ppm / year.
But since the calibration standards that I use to calibrate LTZ#2
also drift it will need some years until I know the whole truth.
...
With best regards
Andreas
I hope at least getting 2 new 8.5 digit measurement points on the diagrams 8)
..Will we three be able to check that coming Saturday?
Frank
Hi Frank,
I'm with you on saturday, together with Keithley 2002 (last Cal in 11/2015) and my Prema 5017 SC (not Cal'ed yet). What else I come up with, well, let's see.
Klingt als hättet Ihr was vor! :)
So you want to use SMD resistors?
As you can read in this thread I've finally used a pickaback board with the gain setting SMD resistors (5ppm/K) and a crystal heater on top of them, to avoid temperature changes. Seems to work quite well.
However, I would spend at least some buffer amp for the LMx99.
I would thermally decouple the 85 deg LM399 from the room temperature NTC and AD587.
You have anyway at minimum 2-3 deg C temperature difference between NTC and the chip of the AD587.
The AD587 has a bad PSRR. (10ppm/V) So I would not put the 15V stabilisation too far away from the cirquit.
Will be interesting to see how the result for the LM399+AD587 combination will work.I'm also a little bit curios...
I have thought of testing this but with the SVR-T I think some of the benefits vanished.Indeed! My breadboarded and not perfectly compensated unit runs stable since 7 weeks, varying only 15-20µV over the changing of room temperature.
I think Joe did the combination to get away with the worse temperature coefficient of the AD587LQ compared to the LM399.Yes, but i think, most of the TC is produced by the resistors and the amp and not by the unheated zener.
Indeed! My breadboarded and not perfectly compensated unit runs stable since 7 weeks, varying only 15-20µV over the changing of room temperature.That could easily be variations in the DMM if you haven't compensated for it
I am missing the cirquit diagram for your PCB.
One problem might be to set the current stable (1mA within 0.1%) through the LM399 zener.All resistors are 0,1% with 25ppm.
The AD587 has a 4K source resistance at the NR pin.The internal source will be nearly total suppressed by the LM399.
Would be interesting how different AD587 behave for the 2 halves of cirquit.I bought several LM399 and AD587, but they are still on their way to my work bench.
I would have used less current here, just to keep selfheatung lower - but thats a minor detail.
You should also consider paying Joe Geller a small royalty fee for each of these circuits you build-- (it *is* patented after all)... I don't know how much he would want, but if you get in touch with him you can negotiate a number...
I also liked that Geller Labs was so open with design, schematic and BOM´s (including my NTC compensation with both first and second order compensation that Joe could have kept for himself)
At first, I thought like you did-- if not for commercial use, then no fees, right? Well, I was wrong. Evidently, it's OK to experiment [research] with patented ideas, but as soon as you put them into actual use, then you are benefiting from the patented design, and the owner of the patent is entitled to compensation.
The substrate is electrically connected to the negative terminal
of the temperature stabilizer. The voltage that can be applied to either
terminal of the reference is 40V more positive or 0.1V more negative than
the substrate.
how much current does the cirquit take?
(if the heater is off then the cirquit is relative instable).
What is extreme noisy (uVpp)? Which Frequency range did you measure?
Power supply rejection should be below 2 ppm/V above 15 V heater voltage.
55ppm voltage difference in a 1Vin change (10.5V to 9.5V). :--
I think the magic smoke is out of this vintage 1971 lm199.
On the other side 20mA is at the lower limit of a LM399.I'm actually testing some LM399H. The older ones (NOS) from 1989 take about 28mA @15V at room temperature, the newer ones (used) from 1998 are running at 35mA.
My LM399 take usually 25-30mA at room temperature at 9.35V with some additional isolation.
is there a correlation of zener voltage to tempco on your references?Yes, like your's and i wonder about the much higher deviation from cold (18°C + 2K self heating) to hot of mine.
the "sweet spot" seems to be at 6875 mV.Would be nice, to get pre-selected parts from some ebay sellers ::)
Would be nice, to get pre-selected parts from some ebay sellers ::)
In my dream world the manufacturer sends all his production of LM399s to his primary precision instrument manufacturer.
.....
The rest is then sent back to the IC manufacturer and is stamped with LM399.
I have put several measurements into one single graph.
In my dream world the manufacturer sends all his production of LM399s to his primary precision instrument manufacturer.
.....
The rest is then sent back to the IC manufacturer and is stamped with LM399.
That sounds too bad, to be false...
So either the process is not well controlled (sounds not plausible to me)
or someone did some cherry picking...
It seems, we have to look out for parts with the magic datecode... 8)
Perhaps better to bodge up test jig with SMU to ramp zener current from 100uA to 10mA in 100uA steps and with Heater on/off switch?
Thought I'd share my build of the simple LM399 based transfer.
Perhaps better to bodge up test jig with SMU to ramp zener current from 100uA to 10mA in 100uA steps and with Heater on/off switch?what do you want to do with that?
below 0.5mA the behaviour is undefined (according to datasheet).
above that you should measure the differential resistance of the zener (around 0.5 Ohms).
https://www.taobao.com/view_image.php?spm=a312a.7728556.2015080703.1.7UZkf9&pic=HEYVDllCEghRV1dcWwocCUQOFx0XTEJfW1lXDBZFAlpcMAl9eXE7MRRUDzQuE1BhaERiZwNbQlpeXV0OSFM=&title=TE0zOTkgUENC&version=2&c=MWIzOTMwMTNhMzgzMDI0OWY4MzY3ZGNiNjE5OWM4NDA%3D&itemId=14154303091&shopId=58559233&sellerRate=25555&fv=9 (https://www.taobao.com/view_image.php?spm=a312a.7728556.2015080703.1.7UZkf9&pic=HEYVDllCEghRV1dcWwocCUQOFx0XTEJfW1lXDBZFAlpcMAl9eXE7MRRUDzQuE1BhaERiZwNbQlpeXV0OSFM=&title=TE0zOTkgUENC&version=2&c=MWIzOTMwMTNhMzgzMDI0OWY4MzY3ZGNiNjE5OWM4NDA%3D&itemId=14154303091&shopId=58559233&sellerRate=25555&fv=9)
...bad link...
What is this PCB?
LCD opening was made by CNC milling machine. All others - by hand drill.
Hi, :)
Kridri,
Is this wat your looking for?
www.bramcam.nl/Diversen/EDN-Split-a-temperature-degree-to-10uC.pdf (http://www.bramcam.nl/Diversen/EDN-Split-a-temperature-degree-to-10uC.pdf)
Kind regarts,
Bram
@ kridri : I have looked for these 2 articles too, I was never able to find them. Especially nothing on Worldcat, it means that unfortunatelly there's no copies of these papers publicy available.
In google scholar, if you type the title and include citations in the search results, you can discover that most of citing papers were written by Jim Williams himself. The others were written in the 70's so there no chance that you can get a copy by contacting the authors.
https://scholar.google.fr/scholar?cites=9898204615121267227&as_sdt=2005&sciodt=0,5&hl=fr
I thought about contacting the MIT library, but as there's nothing in their catalog about this paper, I doubt they could do anything. Maybe the best chance is to contact the Linear Technology staff? I'm quite sure Jim would have kept a copy of this paper in his library.
BTW, if you're looking for papers about sub-milidegree temperature control, plenty were published in the 70's in the scientific journals. BTW-bis, another companion-paper for the MIT oven : "McDerrnott, James. 'Test System at MIT Controls Temperature of Microdegrees." Electronic Design (January 6,1972)" (Not available online).
Laser printer film and 3M adhesive :)
Laser printer film and 3M adhesive :)
I like to do the same thing. Print reversed on an overhead transparency sheet, spray with 3M 77 adhesive, flip over, and apply. Then the Mylar sheet protects the printing.
Schematic is same as on original LM399 boards I had, but layout was slightly revised. Both LMx99's are to be assembled on top side now.
Hello,
how long I need to aging this LM399 with lets say edwin resistor before I can calibrated it against "real volt standard"
Just another DIY poor man's selfcal DVM & voltage/resistance calibrator :) ADS1256+AD5971B
1 ppm INL, 0.25 ppm p-p noise (DCV output), 3 ppm INL (DCV input).
Just another DIY poor man's selfcal DVM & voltage/resistance calibrator :) ADS1256+AD5971B
1 ppm INL, 0.25 ppm p-p noise (DCV output), 3 ppm INL (DCV input).
but where did you hide the LM399.
In one of the black boxes near the statistical divider?
With best regards
Andreas
Quote: "Just another DIY poor man's selfcal DVM & voltage/resistance calibrator :) ADS1256+AD5971B"
Part number for DAC is AD5791B, and is found here:http://www.analog.com/en/products/digital-to-analog-converters/da-converters/ad5791.html (http://www.analog.com/en/products/digital-to-analog-converters/da-converters/ad5791.html)
For those of us that like to play along at home. 8)
Just another DIY poor man's selfcal DVM & voltage/resistance calibrator :)
You do realize that anything past the 6th decimal place in those voltage readings are essentially garbage, no accuracy to speak of? You realize those calculated amounts you posted are also mostly garbage numbers, assuming that the figures used to calculate them are good to the 6th decimal place, that puts the limit of the calculations well short of the 13 decimal places you display and the std. calculated value isn't really any good past the 2nd or 3rd decimal so why even bother putting them there? Really, sloppy work as Bob Pease would say.
The 399's can certainly be pretty good performers even though they can be fairly noisy. Nice devices for 6 digit meters and devices that need power cycling - otherwise their relative high noise can be a problem.
The 399's can certainly be pretty good performers even though they can be fairly noisy. Nice devices for 6 digit meters and devices that need power cycling - otherwise their relative high noise can be a problem.
Ok that seems to be general consensus, noob question: why not L-C filter the heck out of them at the expense of load regulation?
You do realize that anything past the 6th decimal place in those voltage readings are essentially garbage, no accuracy to speak of? You realize those calculated amounts you posted are also mostly garbage numbers, assuming that the figures used to calculate them are good to the 6th decimal place, that puts the limit of the calculations well short of the 13 decimal places you display and the std. calculated value isn't really any good past the 2nd or 3rd decimal so why even bother putting them there? Really, sloppy work as Bob Pease would say.
I am quite sure MisterDiodes was referring to the LTZ compared to the LM399, the LM399 being somewhat more noisier and yes, while you can accomplish some filtering of a reference's output, it must be done very carefully or it can end up degrading the output's performance.
I am quite sure MisterDiodes was referring to the LTZ compared to the LM399, the LM399 being somewhat more noisier and yes, while you can accomplish some filtering of a reference's output, it must be done very carefully or it can end up degrading the output's performance.
The question I was asking really relates to the LTZ or discrete Zeners (and therefore getting seriously OT). I was just curious about the effect that low impedance parallel capacitance might have on the physics of Zener / avalanche breakdown (at the microstructure level) and whether it could cause permanent shifts in the breakdown voltage. Probably of academic relevance only - I just wondered if operating a Zener in the lowest possible capacitance environment could subtly improve its long term drift.
I am quite sure MisterDiodes was referring to the LTZ compared to the LM399, the LM399 being somewhat more noisier and yes, while you can accomplish some filtering of a reference's output, it must be done very carefully or it can end up degrading the output's performance.Sorry - didn't make that clear. LM399 Noise spec is 20uV, 0.5ppm TC, drift is typically lower to mid ppm first year, although can improve with age - they are also tolerant of frequent power cycles. LTZ1000 noise spec is 1.2uV, 0.05ppm TC, drift is typically low ppm per year; usually gets better with time.
Sorry - didn't make that clear. LM399 Noise spec is 20uV, 0.5ppm TC, drift is typically lower to mid ppm first year, although can improve with age - they are also tolerant of frequent power cycles. LTZ1000 noise spec is 1.2uV, 0.05ppm TC, drift is typically low ppm per year; usually gets better with time.
You get what you pay for.
Hmm,
here in the volt nut section we don´t throw away any gear nor any digits >:D
with best regards
Andreas
Sorry - didn't make that clear. LM399 Noise spec is 20uV, 0.5ppm TC, drift is typically lower to mid ppm first year, although can improve with age - they are also tolerant of frequent power cycles. LTZ1000 noise spec is 1.2uV, 0.05ppm TC, drift is typically low ppm per year; usually gets better with time.
You get what you pay for.
Sorry,
but you are comparing apples with pies.
the 20uV(eff) is broadband noise of the LM399. (10Hz - 10kHz).
Whereas the 1.2uV(pp) is 1/f noise (0.1-10Hz).
My LM399 typically are around 4 uVpp (0.1-10Hz).
(some better and some also worse).
with best regards
Andreas
To get the noise down from 4 µV to 1.2 µV , that is a little more than a factor of 3, one needs something like 10 or 12 of the LM399. Reduction of noise goes with the square root.
OH that's right - I had forgotten the older NS datasheet has slightly different specs. It still shows you if you have the opportunity to pick up a working strip chart recorder, they can be very useful at 10Hz! Null meters will tend to have a strip recorder output driver, and that's handy. Plus they make fascinating museum pieces for friends and family to ogle.
Once you hit the sweet spot on your KVD dials, you'll see the meter needle drift slightly and equally around zero point, and you'll get a very accurate idea of your 10Hz and below noise levels.
if noise is your key spec you have to measure every single device.
At least if there are no 100% production tested guaranteed maximum values given (only typical).
Also the noise diagrams seem to be mostly not measured:
how do you explain that a LM329 and a LM399 of LT have exactly the same peaks and dips over time on their low frequency noise diagram.
And how can a LT1021 have exactly the same noise pattern as a LT1236?
Ok they have a similar chip (except the trimming) but how do they get absolutely synchronous noise?
The graphs only represent typical values and for devices which share the same design, the differences are not significant.
By the way: does anyone have a datasheet for the LTFLU-1 with noise diagram?
I have a copy of a NS LM399 datasheet from 1999. It shows noise densitiy curves for 25 C and heated chip. The noise for the cold version only about half the value. The LF noise curve shown is for the heated chip (and different from the one shown below) - looks 8 µV_pp for the 0.01 -1 Hz range.
They also show an internal circuit - identical the the LM329 for the reference part.
So i would expect a lower noise for the relatively cold LM329/LM129 compared to a hot LM399.
The noise density curve for some reason slightly lower for the LTs version of the LM329, but not much.
I struggle to understand why the cold zener chip has less noise compare to hot one?
and second question how to limit the temperature of LM399? What about current limiter for heater?
By the way: does anyone have a datasheet for the LTFLU-1 with noise diagram?There is a datasheet? I surely missed it. :-X
I have tried to get one and gave up.QuoteBy the way: does anyone have a datasheet for the LTFLU-1 with noise diagram?There is a datasheet? I surely missed it. :-X
1. acquire a bunch of LM329's (commercial ones from TI or LT)
2. use some kind of delamination procedure (methods required here!) to strip them off the epoxy/plastic encapsulation.
3. characterize / select these de-laminated parts for an averaging type voltage reference in a custom heating assembly.
4. ie; trying to do an LTZ1000 on the cheap, using selected LM329's.
2. use some kind of delamination procedure (methods required here!) to strip them off the epoxy/plastic encapsulation.
3. characterize / select these de-laminated parts for an averaging type voltage reference in a custom heating assembly.
Just curious- you can still buy all manner of 1Nxxxx series zero TC zeners. I've made some really decent references with those,Hello,
2. use some kind of delamination procedure (methods required here!) to strip them off the epoxy/plastic encapsulation.
3. characterize / select these de-laminated parts for an averaging type voltage reference in a custom heating assembly.
Hello,
I also do not understand why you want to delaminate the epoxy.
I would either use the metal can devices or put the epoxy types into a hermetically sealed housing.
Perhaps with some baking of the epoxy before sealing (with glass).
To reach the LTZ 1/f noise spec you would have to average ~16 LM329 devices (selected for low noise of ~4-5 uVpp)
There was a teardown with pictures of a commercial LMx29 reference here in the board somewhere.
I would start there. perhaps you get the handbook with schematics.
good luck
with best regards
Andreas
Just curious- you can still buy all manner of 1Nxxxx series zero TC zeners. I've made some really decent references with those, an OP-07 and a few resistors, the standard self-biased circuit. Is a 2DW23x really any better than 1N821 or 1N45xx or 1N935-938, or just cheaper?
Just curious- you can still buy all manner of 1Nxxxx series zero TC zeners. I've made some really decent references with those,Hello,
did you measure 1/f noise of the zener?
I got some 1N829A of different manufactures.
NOS of Motorola and ST (cheap) and some new APD (5 EUR / piece).
I measured different noise from device to device (typical 1.7-3uVpp for Mot+APD)
and very different between ST (up to 22uVpp) and the others.
with best regards
Andreas
Hello,
I haven't, but a friend who built commercial equipment did, and he said something interesting. It seems if you select the quietest parts from a batch, those will also have the least long term drift.
Just curious- you can still buy all manner of 1Nxxxx series zero TC zeners. I've made some really decent references with those,Hello,
did you measure 1/f noise of the zener?
I got some 1N829A of different manufactures.
NOS of Motorola and ST (cheap) and some new APD (5 EUR / piece).
I measured different noise from device to device (typical 1.7-3uVpp for Mot+APD)
and very different between ST (up to 22uVpp) and the others.
with best regards
Andreas
I haven't, but a friend who built commercial equipment did, and he said something interesting. It seems if you select the quietest parts from a batch, those will also have the least long term drift.
The zero tc diodes seem to have gotten quite expensive, so maybe selection isn't a good strategy these days. It does make me wonder if one could pot together pairs of conventional (read cheap) zeners that were selected for noise- roll your own!
Hello,
I haven't, but a friend who built commercial equipment did, and he said something interesting. It seems if you select the quietest parts from a batch, those will also have the least long term drift.
thats also something that I read somewhere.
But I still do not know if the 1/f noise or the popcorn noise is meant.
I guess its the popcorn noise since it is related with impurities / imperfections of the silicon.
With best regards
Andreas
i was looking at NZX6V2 which is available @ digikey, and according to its datasheet,
the tempco is +1.5 to 2.5mv/C at 1 to 5ma zener current. Now if this zener family is selected/characterized
for noise performance and paired with a regular silicon diode or a lownoise transistor connected in diode config,
having a tempco of ca. -2.2mv/C, there shall be a "sweet spot" between 1ma and 5ma where these tempco's
shall cancel out completely, and now if properly aged components are used, we *may" have one lownoise (?)
source of steady reference.
Any interesting project with a blown heater lm399? I connected backwards ...
Inviato dal mio Nexus 6P utilizzando Tapatalk
You could test them for temperature coefficient , noise level, and for popcorn noise. Testing for popcorn noise can take a while. Long term drift is more difficult because it is affected by mounting.I second this.
Hi all
I am sorting through some LM399 chips and measuring Vz. Variation so far is 6.86v to 7.15v.
Does this mean anything apart from needing different trimming resistors to get the required 10v buffered output?
enut11
Mimmus,
If you think that lm399 with a blown heater is a better choice for You, You should get hold of an LM329 ( zener half of LM399 no heater ) and run a test with it.
As a comparison
Element14 offer LM399 rated at 2% initial accuracy, and a temp coefficient of 0.3ppm/C at AUD 18.60
LM329 is offered as
Texas instruments product 5% initial accuracy and 50ppm/C temp cpefficient AUD 1.76
and
Linear Technology product unspecified initial accuracy and 15ppm/C temp coefficient AUD 5.00
Even the most rudimentary of ovens should be able to bring in the 50ppm /C specification into check.
I believe every LM399 has their own characteristic when You measure it with 3458A. I have 16 pcs LM399 that already run 24/7 for 1 year even the voltage is "different" the closed one are 3,4mV different, noise is also different. data sheet number is the worst condition and I do not found any exceed the datasheet. so in my opinion if You like to know how good or bad the LM399 with or without heater the best way is compare the healthy one with heater on and then turn the heater off.
surely I like to know how good the LM399 if the heater is turn off when You measure with 3458A.
I think I will find the zero tempco point andzero TC temperature, you mean? From my measurements in the past, that is likely to be quite warm, and most of the ones I measured didn't have a zero slope in the temperature range I put them through.
than use it in some oven project I have in mind. Bob Pease does not recommend to run LM399
without the heater as it leads to instability ...
... or if it is due to the drastical change in weather from yesterday morning (calm) til this morning (stormy with 80km/h) and thus a change in ambient pressure.
The air pressure here at my place in London dropped from approx 1018 mbar on Wednesday midday to 995 mbar by midday today - that's a 2.25% change, not insignificant.
To see the stability of the reference, one should look at the reference voltage only, not the voltage scaled to 10 V. The resistors used for scaling from 7 to 10 V are a second thing, they can drift just as much as the LM399 - this is especially true for the very beginning, when there will be humidity released form the board. A short 12 h test could only identify rare very bad units with maybe excessive popcorn noise.
If at all the slightly higher noise level for the later 2 refs might be relevant, but for the low frequency noise there are also effects like thermal EMF effects and thus temperature fluctuations or air pressure variations that can contribute. So even for the relevant noise the tests might be to short and not well enough thermally shielded.
perhaps you should try to correlate humidity with a PT1-filtered humidity value.
You might want to monitor the '399 output directly, before the booster circuit - just to see if you have a resistor stability issue.
Keep an eye on those SMD divider resistors - they sometimes can pick up FR4 board stresses that show up on the divider ratio output. For precision, on FR4 we'll usually use TH resistors or if we have to use SMT they will go on something like a better stability Rogers pcb material, etc.
wow nearly 7 weeks (continously?) measurement. 8)
how much of the coefficients belong to the LM399 and which part is for the 3458A?
And how much do they add or compensate?
In order to calculate those coefficients, do you need an environment where changes can be isolated?
I performed the compensation for temperature, afterwards for humidity and at least for ambient pressure. Attached are the single steps as a series of diagrams.
The compensation is resulting in:
Mean: 10.001790V
Std: 2.58µV
Is it interpolated to 0 deg C and 0% rH and 0% pressure or what?
It gets not clear to me: you did the correlation one after the other or all coefficients together or incremental (with correction of the previous coefficients)?
On the first view the dew point temperature seems to be better than rH for humidity.
But since the time constants for humidity are in the 3-7 day range a correlation with a pt1 filtered dew point temperature would be also interesting.
.....The effect of humidity is usually not instant. It is not air humidity but more like the humidity contend in the board and parts that influences the circuit. This is delayed against the air humidity. A pt1 filter (= simple first order low pass filter) could be used as a first estimate. However there is still the time constant (e.g. 5 days) that is an additional parameter. Using a filtered humidity is a little like using an intentional very slow sensor.QuoteBut since the time constants for humidity are in the 3-7 day range a correlation with a pt1 filtered dew point temperature would be also interesting.
Still I don't understand what you mean by "pt1 filtered". Maybe you can give me an example?
Still I don't understand what you mean by "pt1 filtered". Maybe you can give me an example?
For me it seems that there is a 1 ppm drift over time additionally to the dew point sensitivity.
It also would be interesting if the result is due to the scaling resistors or the LM399 itself.
For the thermal conductivity it is the absolute humidity or dew point that is important. So 90 % RH at 90°C is far from what one could get from room air. Under normal conditions the partial pressure of water is more like in the 10-20 mbar range, thus something like 1% abs water contend this one would expect a water contribution in the 0.5% range. So it would be a limited effect on the thermal properties.
SMD resistors are sensitive to board stress and thus indirectly may react to humidity. The main effect of a partially sealed case is slowing down changes in humidity. So it can take a long time to get a stable value.
beside amplified popcorn noise I again do have a temperature influence
I did not mean the 0.5ppm popcorn noise but the larger (short) spikes of around 4 ppm especially at the beginning of the measurement.
While the raw reference voltage is almost insensitive to changes in temperature, humidity and pressure, the amplified 10V output indicates a correlation. I'm currently not sure if the resistors haven't yet stabilized in their new environment or if it's the LT1001 which is a CDIP type. But as they are all within a Hammond aluminium die cast and covered with batting I can't find a simple and thus logical explanation. Any ideas on this?
Thermocouple effects?
Did you use the offset voltage adjustment terminals of the LT1001? They will produce an offset voltage drift of 1uV/C for every 300uV of offset away from zero.
QuoteThermocouple effects?
Did you use the offset voltage adjustment terminals of the LT1001? They will produce an offset voltage drift of 1uV/C for every 300uV of offset away from zero.
Well, the LT1001 is in a precision socket, but everything inside the Hammond die cast is filled with batting. I don't use the offset trim pins, they are left open.
I've started measuring the difference between reference voltage and amplified voltage, thus the amp itself. Here are current diagrams. The FFT indicates 1/f-noise as well as the popcorn (burst) noise of the reference. Temperature influence on the gain stage is also clearly visible.
Question here is this. Is the upward slope of this LM399 typical for >500 hours of operation? I doubt the meter is drifting but it is possible I suppose.
While the raw reference voltage is almost insensitive to changes in temperature, humidity and pressure, the amplified 10V output indicates a correlation. I'm currently not sure if the resistors haven't yet stabilized in their new environment or if it's the LT1001 which is a CDIP type. But as they are all within a Hammond aluminium die cast and covered with batting I can't find a simple and thus logical explanation. Any ideas on this?
Thermocouple effects?
Did you use the offset voltage adjustment terminals of the LT1001? They will produce an offset voltage drift of 1uV/C for every 300uV of offset away from zero.
Question here is this. Is the upward slope of this LM399 typical for >500 hours of operation? I doubt the meter is drifting but it is possible I suppose.
Hello,
it really depends. See examples here:
https://www.eevblog.com/forum/metrology/lm399-based-10-v-reference/msg796829/#msg796829 (https://www.eevblog.com/forum/metrology/lm399-based-10-v-reference/msg796829/#msg796829)
So you should wait for at least 5000 hours or 200 days before you can really judge a LM399.
But it could also be the ageing of your resistors. 25ppm/K is not really precision.
And your trimming scheme could also be better (see LM399 data sheet).
with best regards
Andreas
Did you use the offset voltage adjustment terminals of the LT1001? They will produce an offset voltage drift of 1uV/C for every 300uV of offset away from zero.
I did not use the offset terminals in my build. After looking at branadic's data, I should probably see if the amplifier is the source of the drift.
Did you use the offset voltage adjustment terminals of the LT1001? They will produce an offset voltage drift of 1uV/C for every 300uV of offset away from zero.
but I thought the drift would settle down somewhat after about 200 hours like most other references.
QuoteDid you use the offset voltage adjustment terminals of the LT1001? They will produce an offset voltage drift of 1uV/C for every 300uV of offset away from zero.
From where do you get those numbers?
The ratio of 300 µV of offset producing 1 µV/K of drift is a good approximation, but not 100% accurate.
The main reason for offsets in BJT based amplifiers is usually a mismatch in current or effective size of a transistor pair. This results in an offset voltage proportional to absolute temperature (This is one way to make a PAT source) and thus the approximate ratio.
With most BJT based OPs adjusting the offset to zero will also reduce the drift to a low value. However there can be small extra contributions to the drift. The relation between offset and drift is usually better at higher offset.
The min / max values in the datasheet are usually what is tested - so there ratio may not reflect the physics behind it.
One additional contributor to drift in some bipolar OPAs is going to be internal input bias current compensation. The LT1001 has a bias current reduction circuit, so does the OP27 and so does its predecessor the OP07.
In most of these bias current compensation circuits there are two arms to the compensation circuit, one which measures the bias current on one side of the main differential pair (usually from a cascode bias) and one that supplies the 'make up' current to the inputs. They tend to look a awful lot like PTAT current sources.
For low drift applications it might be wise to avoid OPAs with input bias current compensation/reduction circuitry.
One additional contributor to drift in some bipolar OPAs is going to be internal input bias current compensation. The LT1001 has a bias current reduction circuit, so does the OP27 and so does its predecessor the OP07.
In most of these bias current compensation circuits there are two arms to the compensation circuit, one which measures the bias current on one side of the main differential pair (usually from a cascode bias) and one that supplies the 'make up' current to the inputs. They tend to look a awful lot like PTAT current sources.
For low drift applications it might be wise to avoid OPAs with input bias current compensation/reduction circuitry.
This is a deliberate design decision and not a factor. I doubt there is any precision operational amplifier produced which does not operate this way. The internal current biasing of the operational amplifier drives the differential input stage with a PTAT current so that the adjusted input bias current does not vary as much over temperature; it is a deliberate effort at temperature compensation. The input bias current compensation circuit just goes along for the ride.
For those who might be interested in the history of this, the improvement made to the LM301A over the LM301 was to use a PTAT current source to bias the input differential stage instead of a constant current.
I doubt there is any precision operational amplifier produced which does not operate this way. The internal current biasing of the operational amplifier drives the differential input stage with a PTAT current so that the adjusted input bias current does not vary as much over temperature; it is a deliberate effort at temperature compensation. The input bias current compensation circuit just goes along for the ride.
Again, I'm not saying that an input bias current reduction scheme is definitely a source of offset drift, but is something to consider as a possible source of drift.
One thing to watch for on the opamps with a compensated input bias current is the current noise, as it is much higher than that on opamps with a non-compensated input current of the same magnitude.
So what do you suggest as a replacement for LT1001?
There is no free hanging copper tape bridge of special length between the opamp and the resistors to compensate for temperature :-DD
-branadic-
Hello,
again a ageing chart of 2 LM399 references.
now after 2 years (more than 730 days) 24/7 operation. 8)
And probably the last time. Since I may need the multiplexer
channels for two brand new LTZ1000A references soon.
Eventually I modified the circuit to replace the LM1001CN8 with a OPA177FP. Some experiments while building another circuit showed this op-amp superior to the LM1001.
Hello Mickle,
your experiment orientations are that what I call orientation "0" and "5" where I also have only little differences (up to around 0.5ppm) in my setups.
The pin 1 marker directs to the right side on the photo.
So the critical directions would be the putting them on the left and on the right (connector) sides.
I guess that in this case there would be also about 3-4 ppm difference.
When looking at photos from older HP34401a or Keithley 2000 units the critical directions should be putting them on the left and on the right side. The pin 1 marker shows either to the left or the right side in this case.
With best regards
Andreas
Hello,
what has the bad thermal design of the LM399 to do with that weird theory?
with best regards
Andreas
ZENER DIODES QUANTUM DETECTORS DETECTING SPACE DIRECTLYobviously. :palm:
Hello,
what has the bad thermal design of the LM399 to do with that weird theory?
with best regards
Andreas
Well,QuoteZENER DIODES QUANTUM DETECTORS DETECTING SPACE DIRECTLYobviously. :palm:
He haven't even touched on the indirect detecting of space, go figure. :horse:
I think the point was the video claims the orientation of the zeners within the detector and the orientation of the multiple detectors generating correlated noise are what measure the anisotropic "ether" in that presenters theory.
Hello,
what has the bad thermal design of the LM399 to do with that weird theory?
with best regards
Andreas
Well,QuoteZENER DIODES QUANTUM DETECTORS DETECTING SPACE DIRECTLYobviously. :palm:
He haven't even touched on the indirect detecting of space, go figure. :horse:
I think the point was the video claims the orientation of the zeners within the detector and the orientation of the multiple detectors generating correlated noise are what measure the anisotropic "ether" in that presenters theory.
We know that there is no Luminous Aether (Michelson–Morley experiment) ...
Uhmmmm... no. The Michelson-Morley experiment was flawed, but even so, it produced a very small (near the noise floor) positive result. The experiment was later repeated by Dayton Miller (http://en.wikipedia.org/wiki/Dayton_Miller), and using a more sensitive instrument (http://www.orgonelab.org/miller.htm), he obtained very positive proof of an Aether (http://www.anti-relativity.com/daytonmiller.htm). He then died before he could publish, and the university tried to bury his work, but his assistant was able to save enough of it that we now know that he was successful. Shortly after this discovery (or rediscovery), Einstein started to change his attitude towards the existence of an Aether-- and you can read about this in transcripts of various conferences he spoke at. Tesla was right, and Einstein was not exactly correct (http://www.newdawnmagazine.com/articles/tesla-vs-einstein-the-ether-the-birth-of-the-new-physics).
"Experimentum summus judex." (Loosely translated: Experiment trumps theory.)
Note that is was of Mr. Miller's opinion that the Aether is "entrained" by the Earth, and so he conducted his experiment at at high altitude. I doubt that the LIGO experiment is being conducted at high altitude-- "apples and oranges" so they say.
What you just said is what I was eluding to-- that many people have closed minds to alternative theories-- often to the detriment of mankind.
In addition, when we are talking about a theory: "Even in the face of overwhelming positive evidence, if there is one piece of data that is not in agreement with that theory, then that theory is not correct, and you must develop a new theory that better explains the data." -- Richard Feynman
"Experimentum summus judex." (Loosely translated: Experiment trumps theory.)
BTW-- I used to work at E.G.&G. "Special Projects" (Area-51). I've "seen things" that would cause any physicist to doubt their assumptions-- and that's all I can say about that. The Aether is real, it has properties, it's existence and properties are fundamentally important to physics; and the person that manages to figure it all out (and get published) will win a Nobel prize.
What, YOU, worrying?
I've already said enough. I want to keep breathing...
BTW-- I used to work at E.G.&G. "Special Projects" (Area-51). I've "seen things" that would cause any physicist to doubt their assumptions-- and that's all I can say about that. The Aether is real, it has properties, it's existence and properties are fundamentally important to physics; and the person that manages to figure it all out (and get published) will win a Nobel prize.
Oh!! Do tell us more!!!
I've already said enough. I want to keep breathing...
When one of us is receiving our Nobel Prize for Physics for destroying Relativity, he/she will probably have to credit fmaimon for starting a thread on LM399 voltage references. Might have to call it the LM399 10V Reference Theory of Unrelativity.BTW-- I used to work at E.G.&G. "Special Projects" (Area-51). I've "seen things" that would cause any physicist to doubt their assumptions-- and that's all I can say about that. The Aether is real, it has properties, it's existence and properties are fundamentally important to physics; and the person that manages to figure it all out (and get published) will win a Nobel prize.
Oh!! Do tell us more!!!
I've already said enough. I want to keep breathing...
It's too late now, there are some men in black suits, waiting for you downstairs ... :)
Hello,
The circuit shown here: http://www.linear.com/product/LM399 (http://www.linear.com/product/LM399) apears seems simple, some recommendation to build or "improve" stability? Is missing something?
(http://cds.linear.com/image/4908.png)
Thanks!!
It can be improved a bit by getting the current to drive the LM399 to a large part from the 10 V output instead of the 15 V supply. Another point worth adding is a RC filter to reduce the higher frequency noise of the LM399.
Heater must use separate wires that go to PSU.
You need kelvin connection to zener output too.
Yea, something like that. I'd go for additional tap (thin wires is ok) for voltage sense. So you have 6 wires going out of the LM399 ;).
Post the schematic as a PDF or jpg i dont have the cad package you are using to draw itJust do a screen capture. Even with KiCad, you get this as you are using some custom schematic elements:
Hi Rafael, right now, the LM399 zener is using the 15V rail and the 7k5 resistor as a current source. As the current through a zener changes, its output voltage will also change (by a smaller amount). So if your 15V rail drifts a bit, or if the 7k5 resistor drifts a bit, that can affect your output voltage.
There is a zener technique you can use (I believe it is called "bootstrapping"), where the regulated output of the zener is used to drive current through the zener, which means the current through the zener is as stable as the zener itself.
Here's the basic idea: (attached). This is the same technique used on the circuit which is on the first page of this thread.
Edit: You may also be interested in some of the posts in this thread: https://www.eevblog.com/forum/metrology/influence-of-resistors-in-lm399-reference-circuit/ (https://www.eevblog.com/forum/metrology/influence-of-resistors-in-lm399-reference-circuit/)
Best to use a PDF for schematics, or a PNG for graphics / charts you want to be seen embedded inline with your posts. JPG is great for photos but introduces artifacts due to its lossy compression which can be horrible for diagram type drawings.Post the schematic as a PDF or jpg i dont have the cad package you are using to draw itJust do a screen capture. Even with KiCad, you get this as you are using some custom schematic elements:
What, YOU, worrying?
I've already said enough. I want to keep breathing...
Indeed, DiligentMinds is now gone--I hope he *is* still breathing.
DiligentMinds, wherever you are, we miss you.
It's a great hobby to draw pcbs.... ;D
It's a great hobby to draw pcbs.... ;D
And one can do so many re-designs:
I fear that the heater current (change) will influence your output voltage.
Why do you refer the ground pin of the 20K resistor to the input and not to the output?
with best regards
Andreas
interesting oxidation of LM's legs, it actually from ebay ( seen as very beginning of leg are OK )
around 6 month under +160 C; 1-st step done, will set a drifting measurement now ...
under board a thick thing are copper wire; board a green pcb, resistors blue :)
interesting oxidation of LM's legs, it actually from ebay ( seen as very beginning of leg are OK )
(https://preview.ibb.co/bL4dYc/DSCN1862.jpg)
ОК, question:
I hook 5 of that burned thingi ( LM399), feed from precision V source , toss in thick Styrofoam box, and measure Vout precisely ( in use HP3456A; temp env. +- 0.3C ) ...
that was APR 28, now second measure today - MAY 06 ( 8x24 approx 200 hours ) - I got identical numbers, for all 5.
Normally, how long I should wait to get at least one last digit ( 5-th digits after comma) change for LM399.
It's a great hobby to draw pcbs.... ;D
And one can do so many re-designs:
I fear that the heater current (change) will influence your output voltage.
Why do you refer the ground pin of the 20K resistor to the input and not to the output?
with best regards
Andreas
Thanks for the tip...
I will update.
just an update:
may 29 -ON june 24 - OFF for LM drifting measure:
in ppm, in the end of period:
1: -2
2: -47 (weird jump)
3: -3
4: -2
5: -5
6: -1
7: -2
8: i blew it up ...
it can be multi-meter drift, i use only one unit (3456) to measure periodically turn it on every week, and env temp very stable +18.7 (basement). at least 4 candidates ... As i assumed, do it in parallel doesn't increase the precision, as all of then will drift in the same direction with similar speed .....
Four years ago Bob Dobkin said at "Diligent Mind":
"The LM399's do not age when they are turned off and have almost no hysteresis-- so keeping the reference *off* until a few hours before you need to use it [and/or calibrate it] is the best way to keep the long-term drift minimised."
Cheers
I do not know the truth, but now I quote the whole sentence as from «Reply # 635 on: March 13, 2014, 08:02:10 AM»(now deleted):
6) After telling him that I wanted a voltage transfer device, he said a better way might be to use [at least 6] LM399's in parallel [like the Bob Pease idea]. He said that the burn-in procedure would be to operate these in an oven set to 125C for 2 weeks, which would be equivalent to 1000's of hours of normal operation. Any LM399's that are drifting too much after that can be replaced [i.e., you burn-in more than you need, and select the best units for the array]. The LM399 is much more sensitive to board stress than the LTZ [because the LTZ has a special mechanical arrangement in the die mount]-- so the LM399 should be mounted off of the PCB a little bit to allow for this. The long term stability of the array of LM399's will be directly related to the power required to run the heater-- and this can be minimized with insulation-- the more the better! The LM399's do not age when they are turned off, and have almost no hysteresis-- so keeping the reference *off* until a few hours before you need to use it [and/or calibrate it] is the best way to keep the long-term drift minimized.
7) Note that for LM399-based designs, the slots in the PC board [plus a lot of insulation top and bottom] make sense-- Bob said that the less power the heater requires, then the more stable the output voltage will be. So, in this case, the slots [plus insulation] are helping with this. Oh-- and he also said that the LM399 should be run at about 1mA of Zener current for best stability. The more stable you can make the Zener current, the more stable will be the output voltage. He said that there is about 1uV of voltage change for 1uA of current change.
Quote: " ... datasheet for the LM399 suggests a 7.5K resistor with a 15V supply, which gives 2mA ..."
That's an old trap I guess we have all gone into here or there or somewhere else: 15 Volt over 7.5 kOhm is 2 mA, but 15 Volt minus the 7.x Volt over the LM399 gives - Yes, a tad more than 1 mA through 7.5 kOhm.
The HP in the HP3456 multimeter uses a current of 1.3mA - 1.4mA.
Too bad that as isolation use only the cup made of valox and it is known that moving it varies the voltage probably because it varies the thermal gradient
Since temperature and resistance matching is going to best for the resistors in the same package, perhaps parallel two packages. For example, for the top half of the ratio, use 1 from package A and 2 from package B. Then for the bottom half of the ratio, use 7 from package A and 2 from package B. That produces a 3/7 ratio such that only the ratio within each package needs to be maintained. The combination is 2.2 times the value of a single resistor, so maybe the 5k resistors makes more sense. But with so many ratios available this way, one could choose something closer to what their LM399 actually needs to minimize trimming.That is an interesting arrangement of resistors, but im not sure if it works the way you described. To simplify the mental math, lets grossly exaggerate the problem. Lets say the resistance of one of the resistor networks increases by few orders or magnitudes, the overall ratio of the arrangement would then be dominated by the other resistor network. A quick calculation in excel shows that a 5ppm change in one of the networks will lead to a ~1.2ppm change in my circuit, and ~2.2ppm change in the arrangement you've shown. Can you confirm my math?
I see six wires going to the LM399 so Kelvin connection. But I'm not sure where exactly those would go. One 0V wire to your output terminal and another to the 0V of the op-amp, I guess. And one 7V wire to the op-amp input and another to the resistor supplying the zener current?
IMHO the LM317 is a lot noisier than even more classic regulators like the LM723 with a PNP pass transistor.
DK7JB has created a paper (unfortunately in German) comparing a lot of voltage regulators for output noise (just look at the graphs). The LM723 won hands down.
https://www.bartelsos.de/dk7jb.php/rauschen-von-spannungsreglern?download=115 (https://www.bartelsos.de/dk7jb.php/rauschen-von-spannungsreglern?download=115)
IMHO the LM317 is a lot noisier than even more classic regulators like the LM723 with a PNP pass transistor.
DK7JB has created a paper (unfortunately in German) comparing a lot of voltage regulators for output noise (just look at the graphs). The LM723 won hands down.
https://www.bartelsos.de/dk7jb.php/rauschen-von-spannungsreglern?download=115 (https://www.bartelsos.de/dk7jb.php/rauschen-von-spannungsreglern?download=115)
All of the integrated regulators which use a bandgap reference are pretty poor compared to the zener reference on the 723. They not only have have higher reference noise but must multiply it to get their output voltage.
I would like to see how using a 317 as the pass element of a 723 would perform. Enclosing the noisy 317 inside the 723's feedback loop and low noise reference should work pretty well. The advantage is integrated protection of the pass element.
I have never seen this arrangement, but I could give it a try when I'm back at the bench. While protection is cool, I am afraid that the 317 produces more noise than the LM723 is able to kill (the error amp gain is only 33dB).
If you want a protected pass element, then the LM395 would be a better choice. No bandgap reference at all.
Adding current limit to the precision regulating loop adds noise.
Powering a LM399 should be done by a supply with minimum noise.
Actually the PWM is only to boost the zener output voltage to 10V. I'm still waiting for the boards to arrive. Customs took quite long as the boards arrived in Frankfurt on 22th of november, so 10 days ago.
-branadic-
Expanded graph sections from the results above. All measurements are spot checks.
The blue line is Vz. Graph shows variation from test-to-test.
The red line is 10v output. Graph shows variation from 10v. I occasionally tweaked the 10v output to bring it back to 10.00000v.
Some of the graphs show off-chart excursions. These are simply missing data from when the Refs were traveling around Australia for the Aussie Cal Club.
I am zeroing in on Ref #5 as the best of this lot.
enut11
It's possible to do the ratio with one package, as 1.5k/3.5k.
These packages contain eight 25ppm/C resistors, tracking to 5ppm/C. Note that the datasheet specifications are given as "typical".Diligentminds had an interesting post about these, saying that after initial burn-in, they are much more stable.
Individual resistor values of interest include 1k 2k, 5k, and 10k.
Then z01z suggestedYes, then I realized that for having a statistical divider, it is better to have more parts. For example, the single R in the upper resistor is the most sensitive to changes.
Quote from: z01z on June 14, 2018, 05:38:26 pm (https://www.eevblog.com/forum/index.php?topic=15982.msg1607323#msg1607323)It's possible to do the ratio with one package, as 1.5k/3.5k.
Yes indeed, and surprisingly (to me at least), the 10/7 ratio is independent of the individual resistor values.This depends on how ideal your opamp is, you wouldn't get away with say, using a 1Ohm or 1GOhm resistor.
JimmyJo built a reference with a 7V to 10V gain stage using a Vishay NOMCA resistor array
https://www.eevblog.com/forum/metrology/lm399-based-10-v-reference/msg1604887/#msg1604887 (https://www.eevblog.com/forum/metrology/lm399-based-10-v-reference/msg1604887/#msg1604887).
These packages contain eight 25ppm/C resistors, tracking to 5ppm/C. Note that the datasheet specifications are given as "typical".
Individual resistor values of interest include 1k 2k, 5k, and 10k.
Then z01z suggested
Quote from: z01z on June 14, 2018, 05:38:26 pm (https://www.eevblog.com/forum/index.php?topic=15982.msg1607323#msg1607323)It's possible to do the ratio with one package, as 1.5k/3.5k.
Yes indeed, and surprisingly (to me at least), the 10/7 ratio is independent of the individual resistor values.
(https://www.eevblog.com/forum/metrology/lm399-based-10-v-reference/?action=dlattach;attach=589294)
Has anyone calculated an overall temperature coefficient for this circuit?
After some progress the "ION" is up and running!
It needs some care but should not be so hard.
It needs some care but should not be so hard.
Lets do the math for the current cirquit.
LT1763 has up to 60mV change in bandgap voltage (1220 mV) over a 165 K temperature range.
so worst case up to 300 ppm/K.
Personally I'd have used jellybean opamps and pnp transistors to achieve similar noise (why don't they specify <10 Hz noise for many/most voltage regulators?).
the overall TC would not be much worse than the TC of the LM399.
Obviously a good idea.
I think we should take this modification into the re-design.
If no actively enabled the AVR internal pull ups should be off.
Saw during setting the fuses there is a ATTINY15 compatible 6.4 MHz mode
With this clock we are at 5.5 mA at 5V already.
But something is wrong as the PWM clock drops to 800 Hz.
Hello,
now I fear I have done too many modifications in one step:
low noise modification according to Kleinstein (R15 tap)
low power modifications to 6.6 mA
- R8 = 100K
- 8 MHz internal R/C oscillator
- sleep mode active
T.C. increased by a factor of 10
The whole measurement is also relative instable.
there are around 2 ppm changes without any large environment changes.
(perhaps some EMI related influence).
I decided to revert the low noise R15 tap for the next T.C. measurement.
next measurement will be with 4.9152 MHz XTAL which looks promising with ~4.5 mA.
AFAIR the ISR was more like 20 cycles.
Can you check wether you have the same instabilities with the R/C clock as I have here?
Could also be that I got a bad batch of ATTINies on my side.
/*******************************************
Original code: adver ([url]http://radiokot.ru/forum/viewtopic.php?p=3112844#p3112844[/url])
Modified & ported to AVR Studio: Mickle T.
*******************************************
Compiler : WinAVR
System Clock : 16-20 MHz
Fuses : Ceramic Resonator, NO prescaler
*******************************************/
#include <avr/io.h>
#include <util/delay.h>
#include <avr/interrupt.h>
#include <avr/eeprom.h>
//******************************************
#define BASIS 14 // 14 bits of fractional part (8+14bit total => 2.4uV resolution)
#define DIVISOR (1<<14)
#define Key_Up() !(PINB & ~(1<<PB1)) // Is "UP" pressed?
#define Key_Down() !(PINB & ~(1<<PB2)) // Is "DOWN" pressed?
//******************************************
unsigned long Value = 2885765UL;
// Value have a fixed point format [8.14]
// 256/Value = amplification factor of 7-10 V converter
// If LM399 output voltage = U_Zener, then
// Value = (U_Zener/10.000 V)*256*(1<<BASIS) = (U_Zener/10.000 V)*4194304
// Value = (6,877576V / 10V) *256 * 16384 = 2884664,4527104 = 2884664
//******************************************
// MASH internal variables
unsigned int ValInt; // Integer part of Value
unsigned int ValFrac; // Fractional part of Value
unsigned int q[3]={1,0,0}; // Latch outputs 16bit
char c[3][3]={{0,0,0}, {0,0,0}, {0,0,0}}; // Carry flip flops 1bit
unsigned int d[3]; // Adder output / latch data input 16bit
//******************************************
// Calibration factor (Value), saved in EEPROM
EEMEM unsigned long CFG_CAL_VALUE = 2885765UL;
//******************************************
void Init(void)
{
DDRB |= (1 << PB0); // PWM active
DDRB &= ~((1<<PB1) | (1<<PB2)); // UP & DOWN key pins config
PORTB |= (1 << PB1) | (1 << PB2); // Pullup resistors
ValInt = Value >> BASIS;
ValFrac = Value % DIVISOR;
OCR0A = ValInt-1; // PWM compare match register initial value
TCCR0A = (1<<COM0A1) | (1<<COM0A0) | (0<<COM0B1) | (0<<COM0B0) | (1<<WGM01) | (1<<WGM00);
// Set OC0A on Compare Match, BOTTOM, Normal mode, OC0B disconnected, Fast PWM Mode (WGM = 3)
TCCR0B = (0<<WGM02) | (0<<CS02) | (1<<CS01) | (0<<CS00); // Prescaler CK/8
TIMSK = (1 << TOIE0); // Enable overflow interrupt
sei();
}
//******************************************
ISR (TIM0_OVF_vect)
{
// Carry flip-flops
c[1][1] = c[1][0]; c[2][2] = c[2][1]; c[2][1] = c[2][0];
// Adders
d[0] = (q[0]+ValFrac) % DIVISOR;
d[1] = (q[1]+d[0]) % DIVISOR;
d[2] = (q[2]+d[1]) % DIVISOR;
// Carries
c[0][0] = (q[0]+ValFrac) >> BASIS;
c[1][0] = (q[1]+d[0]) >> BASIS;
c[2][0] = (q[2]+d[1]) >> BASIS;
// Latch
for (int i=0; i<3; i++) q[i] = d[i];
OCR0A = ValInt - 1
+ c[0][0]
+ c[1][0] - c[1][1]
+ c[2][0] - 2*c[2][1] + c[2][2];
}
//******************************************
// PWM configuration update
//******************************************
void Update(void)
{
cli();
ValInt = Value >> BASIS;
ValFrac = Value % DIVISOR;
OCR0A = ValInt-1;
sei();
}
//******************************************
// Loading the calibration factor (Value) from EEPROM
//******************************************
void ConfigLoad (void)
{
// If the both keys was pressed on the startup,
// only the failsafe initial value is loaded !!!
if (!Key_Up() && !Key_Down()) {
eeprom_read_block(&Value,&CFG_CAL_VALUE,4);
_delay_ms(3000); };
}
//******************************************
// Update calibration factor in EEPROM
//******************************************
void ConfigUpdate (void)
{
eeprom_update_block(&Value,&CFG_CAL_VALUE,4);
_delay_ms(3000);
}
//******************************************
int main( void )
{
ConfigLoad(); // If UP & DOWN keys was pressed at startup
// then load default config
Init();
while(1) {
if (Key_Up() && Key_Down()) // If UP & DOWN keys was pressed
ConfigUpdate(); // then save config to EEPROM
if (Key_Up() && !Key_Down()) {
Value ++;
Update(); };
if (!Key_Up() && Key_Down()) {
Value --;
Update(); };
_delay_ms(300);
};
}
I used my scope for this since I do not have a frequency counter.
With the 20 MHz XTAL we get 0.002 Hz standard deviation.
The 8 MHz R/C oscillator shows 4.3 Hz. So that explains why I have such a instable 10V output with the R/C oscillator.
The 4.9152 MHz XTAL oscillator has 0.00044 Hz standard deviation. (so around a Factor 4 lower than the 20 MHz)
I guess that I had a temperature without excessive noise for the ~5 MHz XTAL.
because the work you do is amazing.
So relative to their frequency the jitter is the same for both crystals (20MHz: 0.1ppb, 4.9152MHz: 0.09ppb) while the R/C oscillator shows 0.5ppm jitter.
Would be interesting if the R/C oscillator shows a much more jitter also on your side.
For exact comparison, how did you set the fuses for use of internal R/C oscillator?
I like this research here very much. It shows that many things aren't that easy in pratice. Starting with the RC oscillator and internal xtal oscillator also shows a lot.
So to meet your requirements you need a crystal of low frequency and good temperature stability. What do you think of using Euroquartz 6,0 MHz MQ?
EDIT: You could also turn of 10V boost circuit during transport to save another few mili amps.
It is more the short term fluctuations of the combination of XTAL and Oscillator cirquit at certain temperatures.
Hello branadic,
unfortunately there is only a ratio 1:1 or 1:8 as pre-scaler for the used PWM-unit.
The interrupt routine takes around 300 clock cycles (at the 22 bit version of the SW).
(so with 1:1 the interrupt routine would calculate longer than the PWM cycle)
So the PWM-frequency has to drop to 2400 Hz.
I have some strange effects with ~5 MHz now where I do not know if I have contact problems or if the ATTINY oscillator cirquit has generally problems with lower frequencies. -> will have to repeat some tests.
Also the 8 MHz R/C oscillator shows instabilities (high T.C.) here as already shown.
Can you check wether you have the same instabilities with the R/C clock as I have here?
Could also be that I got a bad batch of ATTINies on my side.
with best regards
Andreas
Also, what about ARM (Cortex M0 for example) ? Personally I haven't used them before, but I believe this may be much more efficient with integer arithmetic and will definitely have several 16-bit PWM.FYI - these are the 16bit HW PWM module capabilities with STM32F103 @72MHz clock (PLL). That PWM performance is typical for all 16bit HW PWM you may find in those 32bitters.
I have obviously a hardware problem (with the stability of the oscillator) and every body is discussing about software or "better 16-32 bit processor (at best with PLL)" or making wild assumptions on transportation modes which I never intended.
I think its time to do some stability measurements with the new filter caps.
with best regards
Andreas
Regarding the ripple, what if, as a test/temporary solution, instead of lowering MCU consumption you try a reverse way - don't allow MCU to rest ?
This way instead of sleep mode in the main loop, this will become busy loop, doing anything (maybe just the same detection of button press).
The new code looks odd - very much like the PIC code, but I don't see how the information flow from the q to the c values should work. I only see the shifts for the c's and no effect from the q's. So I guess there is something wrong in the code.
Second: Andreas, were you able to verify the simulated results by real world measurements in time and frequency domain? Up to know I couldn't measure such voltage swings.
Third: What if you change your simulation to R15 tap? Any improvements?
-branadic-
The new code looks odd - very much like the PIC code, but I don't see how the information flow from the q to the c values should work. I only see the shifts for the c's and no effect from the q's. So I guess there is something wrong in the code.
You have to think in assembler.
It is essentially the "Bresenham" code but 3rd order interleaved.
with the 8.14 code I have to shift all fractional values by 2 bits left (ValFrac4 = ValFrac * 4)
So I get an overflow into the C bit of the Status register when 65536 or above is reached.
The inline code shifts this Carry-Bit directly into the Bit 0 of the C0, C1 or C2 variable.
All other bits are shifted left (so I avoid the assignments from c[2] = c[1] ...)
In the disassembler the ROL command is translated as ADC Rx,Rx (which is the equivalent).
I have tested both codes in simulator in parallel and have compared the output values over
a simulation time of > 70 seconds real time. (so more than 10 cycles of the random code).
the error counter of comparison of the two results was zero the whole time.
So dont tell me that it does not work.
With best regards
Andreas
The extra oscillation (which should better be called ripple) is likely due to charge injection from the CMOS switch.
If one is after lower ripple, one could consider having the first filter cap (C10/ C3) no towards the OPs output but to ground.
I've a different value for C6 in your LTSpice, which I changed to 100nF also on my board.
Attached is the circuit for cross checking. You can see that with R15 tap (vout2) there is no oscillation, while without R15 tap (vout1) it's clearly visible.
-branadic-
Edit 3
79 kHz
8+14 bit
source code
- R15 tap modification should be implemented in the board (did that already with bodge wires)
- C6 (simulation) / C22 (board files) should be changed from 10nF to 100nF (did that already on my board long time ago)
- replace SMD crystal by HC49 as this gives more flexibility in choosing the frequency value (e.g. from 2.5MHz up to 20MHz, while 2.5MHz has the advantage to surrender 1/8 prescaler and reduce power consuption of the circuit, while PWM frequency remains the same 10kHz)
- additional 100nF MKS2 at the output of second opamp
But, I don't know if the frequency modulation would help or hurt the filtering. Though, it is certainly much more computationally intensive.
Also, how was the 9.6 kHz frequency obtained? Was it because the analog circuit is less ideal at higher frequencies, or due to computational concerns?
@Branadic: can we change C4 up to 10uF in the layout?
I am still missing your source code implementation for the 79 kHz.
I've started working on a PCB design for this project, but with few tweaks. First, I plan to use the STM32G071 CPU, which is fast and pretty low power.
Also, I propose to build a 3-level bit DAC using two synchronized PWM. Since this is geared towards 10V, we can do better if we PWM between 5V and 10V instead of between 0V and 10V. For the 5V output, I will flip-flop between switches (use one for the "low" branch and use the other one for the "low" branch the next cycle). In my current implementation, I've also implemented random frequency variation which greatly spreads out the frequency response, but it may be bad due to offsets caused by unequal rise and fall times of the switch (though this does eventually average out).
I've drawn up an "analog" schematic and I've appreciate feedback.
Things in particular to decide before layout:
- ADG419, ADG1419, or ADG5419. I'm leaning towards ADG1419 due to lower on resistance, but it has larger parasitic capacitances though much lower charge injection
- Switch to LTC2058 (at least for prototyping) since it is cheaper (per op-amp), nearly as good?
- Use ceramic X7R 0.1uF caps for decoupling?
- Add ferrite at switch output (next to R1)??? Perhaps need to build the circuit and try with/without?
- SMD resistors have fewer parasitics, but perhaps we want the series inductance of THT resistors?
- Looking back into the thread, I screwed up the positioning of the 10V node and 100 ohm R4. Fixing that, will upload 20190212b soon.
- Switch to Sallen-Key LPF on the output?
- I want to use the STM32G0 because it's low power, but it seems that the 64-pin package is the best choice. Ugh.
@IMO: nice schematics. From where did you get the model of the ADG419?Bordodynov library. It includes various switches.
The STM32 µC should be able to use 16 bit or similar PWM, which can be a slight advantage with a higher base clock. So less low frequency component. to filter.Above are the tables with stm32 @72MHz and 16bit hw pwm ranges. When not using PLL and targeting "low power" a realistic clock could be something like 24MHz. With 24MHz clock you may get 10kHz PWM with ~11bit resolution.
Switch to Sallen-Key LPF on the output?There is around 800uVpp (@10KHz pwm) ripple at the output of the ADG switch (in my simulation above). You have to decide what is the targeted ripple amplitude at the ION's output. That gives you an idea how to filter it out..
I´m trying different PWM frequencies at the moment.
The frequency has (as expected) a influence on the output voltage in the mV range.
Interestingly the frequency also has a influence on the T.C. of the whole cirquit.
The sweet spot (at least for my sample of ADG419 seems to be between 9.6 and 19.2 kHz (more near the upper frequency)).
Of course this may also depend on if you use 3.3V or 5V input levels.
- Reducing the 100R resistor. Same hope as above since I thougt that the 100R resistor together with the 100K build a voltage divider with the influence lower for lower 100R resistor.
But obviously the ripple is only a thing between discharging C4 over the 100K Resistor R1.
Fine to see a different approach with different ideas. How much current is drawn by your CPU?
I have to admit that I do not understand how your PWM works. (also not familiar with the STM CPU).
Do you have something like a timing diagram?
When combining 2 different PWMs I would have expected a e.g. 100:1 ratio (coarse/fine) between the mixing resistors.
How many bits are used from the PWM?
There is around 800uVpp (@10KHz pwm) ripple at the output of the ADG switch (in my simulation above).
PS: Getting a little bit different results when the V- of the DG switch is connected to -15V instead to the GND.
If the 100R is reduced, perhaps we would run into stability issues due to the large capacitors to ground?
I am attaching a LTSPICE simulation of the dual-PWM scheme. Notice that large spikes on most of the voltage nodes. Not good. As I mentioned, I think they are a combination of charge injection and slow switching times.
attached diagram from measurements of VOut (@25 deg C) and
resulting average (linear) T.C. of my ADG419 switch over PWM frequency. (without R15 tap).
so around + 40.5 uV/kHz near linear change in Vout
and +12.7 ppb/khz change in T.C.
so interpolating: Zero T.C. somewhere near 18.5 kHz
Andreas
I think a better solution would be a good higher order low pass filter. Even the 10Hz 2nd order one in my schematics should filter the 10kHz+ perfectly out.There is around 800uVpp (@10KHz pwm) ripple at the output of the ADG switch (in my simulation above)...you mean the node A2 in your simulation above?
This ripple mainly depends on the 1uF capacitor together with the 100K at the common of the switch (R4/C4) versus PWM frequency.
Thats why I asked branadic to increase the cap to 10uF in our test-board.
Add 10 more phases and get russian V1-18 :-DDShots made by Max Otto von Stierlitz :)
https://www.qrz.ru/schemes/download/4445 (https://www.qrz.ru/schemes/download/4445)
I still see 700uVpp ripple (vref10 node) in the pigrew's simulation.
How should your dual PWM work? What is the trick?
FYI - the best PWM equipped STM32 is the STM32F334 series. It includes 217ps resolution timers.
For example you can generate a 100kHz PWM with ~15.5bit resolution:
144Mx32/100k = 46080 (100% duty)
46080*0.7 = 32256 (70% duty)
For those of you using the ADG419, I'd be interested if you could measure the voltage (versus frequency, constant duty cycle) with the ADG419 vs. the Vishay DG419LE.
Modeled a little ;)
I've realized that the R4 100 ohm resistor is the problem.Not only, also the output impedance of the OP-Amp ... (so reducing the 100R gives no proportional effect)
The ripple is so small I can't zoom in that far in LTSpice.Try V(out) - 10 as formula for the output.
But, there are HF spikes on vx,x1,x2,x3 at the pulse edges.Fortunately they are filtered out easyly by the final low pass filter.
Do we have a target spec for the turn-on transient time?
edit: attached screen shots of spikesThat exactly is the 700-800uVpp ripple I've been referring to (with 1uF cap)..
And you have to filter it out with a quality low-pass filter, to a level which is acceptable by the metrology folk, say, 7nVpp :).
..A low-bias current precision opamp does work, however. In particular the ADA4625-1 is exemplary--it has the triple threat: low bias current, low input current noise and low flicker noise which are all necessary for best results..The guys here are pretty keen on the low voltage offset tempcos, +/-1.2 μV/degC is on the higher side, however.
..A low-bias current precision opamp does work, however. In particular the ADA4625-1 is exemplary--it has the triple threat: low bias current, low input current noise and low flicker noise which are all necessary for best results..The guys here are pretty keen on the low voltage offset tempcos, +/-1.2 μV/degC is on the higher side, however.
I'm getting closer to finalizing a schematic. So, here is my draft 2, with the following features:Would the switches work fine at say 70-100kHz PWM?
- Ability to use my two-phase PWM, or a single phase, ganging together the two SPDT to get a lower on-resistance.
- STM32F334-based PWM, for 250 ps (or so) resolution.
..With the higher PWM resolution, there would be less low frequency part anyway. So I don't think it would need such a high PWM frequency - I would more like expect the best frequency to be lower.Considering the F334's HiRes options the useful PWM freqs are the 8.8kHz, 17.6kHz or 35.1kHz, all w/ 16bits resolution.. 8)
Another point to consider - the HiRes PWM mode of the F334 requires the PLL to be involved. The 217ps and alike resolutions work with 144MHz timer clock only, afaik (doublecheck). With some stm32s (ie the BluePill's F103) the PLL slowly drifts, something like 60Hz @72MHz in a sawtooth manner (not related to sawtooth with GPS). No idea whether that is the case with F334, however. Also the jitter with PLL has to be considered somehow. May be it is not too relevant with 399, but do consider when moving to LTZ1000 :)..With the higher PWM resolution, there would be less low frequency part anyway. So I don't think it would need such a high PWM frequency - I would more like expect the best frequency to be lower.Considering the F334's HiRes options the useful PWM freqs are the 8.8kHz, 17.6kHz or 35.1kHz, all w/ 16bits resolution.. 8)
HRTIM1 timer is made of a digital kernel clocked at 144 MHz followed by delay lines. Delay
lines with closed loop control guarantee a 217 ps resolution whatever the voltage,
temperature or chip-to-chip manufacturing process deviation. (from the datasheet).
Would the switches work fine at say 70-100kHz PWM?
How did you measure that? Did you set the PWM duty to 68% and changed the PWM frequency from 2kHz to 100kHz?Would the switches work fine at say 70-100kHz PWM?My ADG419: no
see T.C. over PWM frequency:
Hello,OK, you kept "PWM duty" constant (hard-coded in your fw) and you changed the "PWM frequency" (also hard-coded in your fw) by replacing the 4 Xtals, and the TC is the lin interpolation between the 4+1 Xtal measurements.
I did 5 T.C. measurements (each lasting one complete day)
every measurement with a different XTAL for the ATTINY85.
(2.45 MHz, 4.91 MHz, 10 MHz, 20 MHz and one with 8:1 prescaler)
The voltage for the overview is measured at 25 deg C.
The T.C. is the linear approximation (slope of the green curves) over ~15 to 40 deg C.
PWM is kept constant. at the precalculated value which should give exactly 10V at the output for the given LM399#3.
with best regards
Andreas
Preparing to tryFYI: Maxim does the similar:NCC-1701Vishay MPM-divider...
It's not exactly 10V output, but cheep and i hope is pretty good stable for my 6.5 DMMs.
..and custom ratios are also available upon request..
..For the output stage the extra transistor could be useful to reduce heating of the OP under heavy load. It would not change much with noise and drift, as the transistor is in the loop.Yes, we talking here the output stage only (399 classic). I don't see any diff against the naked opamp's output in tempco drift. No idea about the added noise, imho it should be by orders lower than the 399. Except the way higher current capability the boosted opamp's output gets some emi/esd protection as well..
Is that an ".io" or the ".ic" (the correct one) LTspice directive in your simulation??
While curious about how your 32bit DAC works and looking at the code and SD DAC whitepaper you refer to, let me kindly ask you - does it mean you modulate (within a periodic ISR) a 16bit PWM duty with the sigma delta modulator's 1bit output?Yes, that's exactly what I do.
Ok, the LSB of the 15.2bit 10kHz PWM "duty" is resolved by the 16.8bit SD modulation, basically. What is the SD's isr() frequency (or period)? [I can see there 3.13us in the comment but it does not fit into my current understanding of the process..]
For the "noise" measurement, I'm getting standard deviations of about 5 uV on a 34401A with 50 samples with NPLC=100 (60Hz). Is this good or bad?
Another idea I had was if it's a good idea to use the LM399 without its heater.
Jitter continues to be an issue. The period of the 100us cycle has a P-P jitter of about 450ns.
There are a number of microcontrollers and digital signal controllers that offer a "High Resolution Timer" in the form of a 16-bit PWM followed by a 32-stage selectable delay line. Some of these control the accuracy of the delay line transparently in hardware, while most of them require firmware support to do this.
As the LM399 is only stable (over the 2 nd year) to maybe 5 ppm, there is not much sense in an adjustment much finer than 20 Bits. So the 22 Bits of the original project is fine.
One point possibly worth a look would be the ADG419 switch. It is relatively slow and might cause some jitter. One might consider a faster switch, maybe even 74HC4053 - it's faster but would need a kind of +-5 V supply.You are welcome to set up a cirquit with the 74HC4053 and make some comparisons.
The '334 hires pwm is with 16bits resolution only. I would be happy to know more chips with 16bit pwm at 4.6GHz pwm clock, indeed.
You can connect an external AND-OR gate (SN74LVC1G0832) to the PWM output and two extra GPIO pins. You set up an interrupt on the positive edge of the PWM and another on the negative edge. These two interrupt routines control a 'PWM-FORCE' signal (which goes to the OR gate input) and a 'PWM-ENABLE' signal (which goes to one of the AND inputs)-- with the original PWM going to the remaining input of the AND gate. This can provide an extra 4-bits of resolution, where the PWM controls both the positive and negative edges (and the force and enable signals are not that critical). This can be (practically) a 25-bit DAC without any dithering at all. If you want even more resolution, you can add (maybe) 3 more bits on the bottom, that dither the 25th bit, providing 28-bits total-- and the code dependent low frequency artifacts will be very small.
Get it now?
You can connect an external AND-OR gate (SN74LVC1G0832) to the PWM output and two extra GPIO pins. You set up an interrupt on the positive edge of the PWM and another on the negative edge. These two interrupt routines control a 'PWM-FORCE' signal (which goes to the OR gate input) and a 'PWM-ENABLE' signal (which goes to one of the AND inputs)-- with the original PWM going to the remaining input of the AND gate. This can provide an extra 4-bits of resolution, where the PWM controls both the positive and negative edges (and the force and enable signals are not that critical). This can be (practically) a 25-bit DAC without any dithering at all. If you want even more resolution, you can add (maybe) 3 more bits on the bottom, that dither the 25th bit, providing 28-bits total-- and the code dependent low frequency artifacts will be very small.
Get it now?
I think I see what you're suggesting.
I think we need to run the final PWM at between 5 and 30 kHz to allow for a reasonable analog filter. The extra bits you output let you construct a few extra MSB that are controlled in software. This also allows the internal timer clock/DLL/whatever to be run at a speed (providing a better time-resolution).
However, I don't see how you could add many more bits without going to much lower resultant PWM frequencies. For example with the STM32F334, the internal PWM period could be up to about 62 kHz (with f_HRTIM=128 MHz*32, 16 bit), and add three extra bits to get a final PWM rate of 7.75 kHz (without dithering). I'm not sure that it's a good idea to go any slower than that.
During my testing of the 'F334, I'm now reading about 53 mA. No matter what I do (other than turning off the HRTIM unit), I can't get the current draw to significantly decrease. I've tried, for example, disabling the unused subtimers and setting their clock dividers to the most divided.
I did also order another dev board today, a TI launchxl-f28027. I'll report on how its PWM generator performs (and current consumption).
A filter for 137Hz is EASY, since you would be filtering the output of the Delta-Sigma integrator, which will have a rather small triangle wave with some spikes due to the finite slew rate of the op-amp. Just a 3-pole filter would do it, and the 137Hz remaining would be well below the noise floor.
What is the -3dB of your 2pole SK filter?
Is the value of R7 ok?
Using Vx to supply the current to the LM399 will only work if the output is a an approximately constant level. If the voltage is adjusted to other levels, it may be better to use a different source (e.g. 15 V supply or maybe another OP for an approximately 9 V).
Speaking of which.... I'm currently biasing the heater with the 15V supply, which seems a bit wasteful of power. How sensitive is the Zener voltage to the heater voltage? Should I bias the heater with the board's input voltage?
(https://www.eevblog.com/forum/projects/lm399-based-10-v-reference/?action=dlattach;attach=44356;image)
Would be great to have some more real data on the pink noise from field.
Would be great to have some more real data on the pink noise from field.
Hello,
there are relative large differences in 1/f noise for the LM399 from sample to sample.
Good devices have around 2-4 uVpp @7V
others i have measured > 6uVpp @ 7V
So if you need low noise you have to sort out the bad ones.
with best regards
Andreas
Hi Andreas,
Have you noticed anything special about the devices with low 1/f noise (Such as lower TempCo, or lower time drift, etc.) ?
-MB
I only need about ten pins, what do I do with the other 80?With the LM399 and the PWM calibrator on your board, you may add 2 opamps (A creating -10V, B integrator), 1 comparator and 3 ADG switches (-10V, 10V, integr cap discharge) and create a 6.5+ digits ADC with all the FPGA's LUTs and IOs available :)
[..]25bits at 1kHz will be a bit of a challenge, but if you could ease the requirements a bit, Parallax' Propeller might be an alternative. That MCU has a quite odd and by now fairly old design, requires support chips (external I2C connected EEPROM), draws quite a bit of power and has other drawbacks. It does however allow for arbitrary length PWM via bit banging (since it has eight 'cogs' -- think of them as cores -- and no interrupts, bit banging can be done jitter-free easily). It's routinely clocked up to 100MHz, some reported success with slightly higher clock rates. Sadly, despite its age, the MCU is notoriously under-specified. It derives its clock from an external crystal via an internal PLL circuit. It's unclear to me how much jitter that introduces (on small time scales -- within a second, I measured the error to be less than 2 parts in 80e6 using a 10MHz reference signal from a GPSDO).
I want about 25 bits of resolution, a 1 kHz PWM frequency, and a delta-sigma loop running at 10 Hz or faster. This implies I want a >167 MHz timer clock (10 Hz * 2^(25-1)). The STM32G01 can nearly achieve that, but I'm predicting its jitter performance isn't so great.
I'm concluding that a FPGA may be the proper solution. The hard-IP PWM/counters I've seen do not have the speed or resolution I want. Restricting my search to non-BGA packages also wipes out much of the competition. The popular ICE40 specifies bad jitter and isn't that fast (I've not tried it). The Lattice MachXO2 seems suitable. Using a "DDR" output, I expect it to be able to achieve an up to 750 MHz equivalent timer clock (twice the internal frequency due to DDR) with the timer logic running at (f/4 = ~90 MHz) (which it can according to static timing analysis). With a slower model in the same line (LCMXO2-7000HE-6 dev board) and an external 80 MHz oscillator, a 500 MHz equivalent timer used to generate a 1 kHz PWM had a jitter of sigma=28 ps, and I suspect this is approaching the noise floor of the oscilloscope (which samples at 4 GSa/s).
[..]
I don't think there is any need to get 25 bits with 1 kHz speed. The LM399 has a much higher noise. So there would be much more noise from the LM399 than any SD modulation amplitude from using "only" some 20 Bit true PWM and than SD modulation on top.
25bits at 1kHz will be a bit of a challenge, but if you could ease the requirements a bit, Parallax' Propeller might be an alternative. That MCU has a quite odd and by now fairly old design, requires support chips (external I2C connected EEPROM), draws quite a bit of power and has other drawbacks. It does however allow for arbitrary length PWM via bit banging (since it has eight 'cogs' -- think of them as cores -- and no interrupts, bit banging can be done jitter-free easily). It's routinely clocked up to 100MHz, some reported success with slightly higher clock rates. Sadly, despite its age, the MCU is notoriously under-specified. It derives its clock from an external crystal via an internal PLL circuit. It's unclear to me how much jitter that introduces (on small time scales -- within a second, I measured the error to be less than 2 parts in 80e6 using a 10MHz reference signal from a GPSDO).
FPGA seems to be the more promising choice, the Propeller might just be easier to program.
I meant to try this for a long time now. You could beat me to it. ;)
Looking at the LM399 datasheet again, it shows ~10 uV (~1.5ppm) of p-p <1Hz noise (Low Frequency Noise Voltage).
As far as I remember noone found a low noise LTC6655 by now. The datasheet values given are misleading as the LNA used to specify noise on them doesn't match the defacto standard bandwidth?The too small bandwidth of AN124 is one reason. The other sad truth is that the LTC6655 noise is very dependant on the input power supply voltage.
Where is the RTN coming from? My guess is the reference, but I'm not sure. I have not done any measurements of Vref. The plot showing RTN is a residual plot, after subtracting out the estimated output voltage based on a linear model of time and temperature.You may try with a "reference" made of a battery cells, like 4x1.5V or 2x3.6V.
The batteries have pretty low noise usually.
Using a FPGA just for the PWM looks like a lot of effort and power consumption.
For the drift / TC I would expect the switches to be a possible source. Some units may behave better than others, as the R_on ratio could be a factor.
[..]I was wondering about that. I don't think max. clock jitter is specified for the P8X32A. Don't really know how to measure it, but recently got a new hammer (LeCroy Wave Ace 2032 which allows fairly fine resolution timing measurements using ETS) ...
The Propeller looks interesting, but probably can't do what I want because of its "only" 80 MHz clock speed (without using an external serializer IC). I'm also worried the low-ish PLL frequency of 128 MHz implies significant clock jitter. [..]
Thanks, I need to ponder this a bit.
Further, the 160ps pk-pk variation in the width of the pulse need to be taken with a grain of salt, as the Wave Ace 2032 specifices:
Trigger and Interpolator Jitter: 0.4ns pk-pk
If you connect the 34401A to a computer anyhow, I would use 10NPLC with 10 samples instead of 100NPLC. The 34401A can't do more than 10NPLC and the math doesn't work correct if the meter does the averaging. And you don't loose resolution above 10V.
I also see far below 1ppm pp with my 34401As
Burn-in of LM399 - do I need to wire the zener reference part as well?Hello,
Does the aging help with the popcorn too?
I've been bending the leads a little bit (using 2.54mm pitch socket). Hopefully that stress does not contribute to the popcorn too..
When using RSx=100k and CLP1=1uF the BW(-3dB) is 1.6Hz.1-pole filter is far from a brick wall filter, so the effective noise bandwidth is 1.57 * 1.6Hz. Not very important here but good to remember. Also watch for 1/f current and voltage noise of the opamps - parameters are specified at 1kHz but you are most concerned about 0-10Hz region. AD app note MT-048 has some relevant information. To be able to ignore resistor 1/f noise make sure to use good quality metal film or wirewound resistors, crappy thick film ones can be problematic.
The OPA2210 precision operational amplifier is built on TI's precision superbeta complementary bipolar semiconductor process, which offers ultra-low flicker noise.. SiGe bipolar process..
[..]I'd appreciate some stated goal, some specifications to be met, to ground the discussion and to evaluate, whether such specifications can (and once realized) have been met. This is even more needed, if a not-so-obvious design is chosen (here: why put the LM399, which has its own heater, in an oven? The LM399 was developed to make ovens or temperature compensation unnecessary).
Shoot @ it! ;)
[..]
the heater (pin 4 on the LM399) might be on a higher potential than low on the "zener" (pin 2 on the LM399).Not if the heater supply is really "floating".
(here: why put the LM399, which has its own heater, in an oven? The LM399 was developed to make ovens or temperature compensation unnecessary).I guess the oven is only for the 7->10V transfer resistors.
Shoot @ it! ;)Id like to hear some of the design decisions / reasons.
Hi,
Maybe I'll start my own topic, but I'm working on this circuit right now.
The circuit will be placed in a small oven that will be kept at about 42 degrees.
The heaters of the LM399AH will get their own "floating" power supply.
And these power supplies will be connected at one point, that is in the schematic by means of R1.
The output buffer is not in the oven but on a separate circuit board.
If there is enough interest, I can create a new topic about this 10V Reference.
The schematic
(http://www.bramcam.nl/Diversen/Chopper-Opamps/LM399AH-10V-Ref-08.png)
Shoot @ it! ;)
Kind regards,
Bram
Does anyone know how I can get the graph in LTSpice so that there are more lines so that it is easy to read?
R5 and R6 are fo charging the capacitor C3 to almost the full reverence voltage, soo over C2 wil than be only a verry low voltage, therefore also very low leakage current.
Temperarture of DUT in °C | Leakage @12.84V in nA | Insulation Resistance in GOhm |
44.7 | 4.12 | 3.11 |
39.5 | 2.45 | 5.22 |
34.3 | 1.49 | 8.59 |
29.1 | 0.89 | 14.38 |
24.0 | 0.55 | 23.27 |
And when I think of what forum user "maat" explained, that the leakage current of this type of capacitor is quite temperature sensitive, it is better not to use this type of capacitor in this project.
@Andreas: my conspiracy theory would be AD/LT is selecting somehow (before a burn-in) the LM399s based on popcorn or 1/f noise level (or Vref<7V?). They sell parts with "LM399" markings to the general public, and the parts marked with "weird numbers" to the selected customers. Guess who gets what..Not quite and reality is not quite as nefarious. There are companies who offer a selection service and it stands to reason that AD/LT is one of those. HP (erm, Keysight) is one of those major customers who pay extra to get units with tighter/guaranteed specifications (and a customized label). Those units are selected from the general pool. The open market gets those which don't quite meet those tighter limits, but also - and that might be the overwhelming majority - all those units which haven't been tested at all. You're free to purchase a set and select the best and sell the rest on Fleebay (which I think some do).
An lawyer would say "KS is not using an LM399H/AH in their DMMs".. :)@Andreas: my conspiracy theory would be AD/LT is selecting somehow (before a burn-in) the LM399s based on popcorn or 1/f noise level (or Vref<7V?). They sell parts with "LM399" markings to the general public, and the parts marked with "weird numbers" to the selected customers. Guess who gets what..Not quite and reality is not quite as nefarious. There are companies who offer a selection service and it stands to reason that AD/LT is one of those. HP (erm, Keysight) is one of those major customers who pay extra to get units with tighter/guaranteed specifications (and a customized label). Those units are selected from the general pool. The open market gets those which don't quite meet those tighter limits, but also - and that might be the overwhelming majority - all those units which haven't been tested at all. You're free to purchase a set and select the best and sell the rest on Fleebay (which I think some do).
@Andreas: my conspiracy theory would be AD/LT is selecting somehow (before a burn-in) the LM399s based on popcorn or 1/f noise level. They sell parts with "LM399" markings to the general public, and the parts marked with "weird numbers" to the selected customers. Guess who gets what..
Not quite and reality is not quite as nefarious. There are companies who offer a selection service and it stands to reason that AD/LT is one of those. HP (erm, Keysight) is one of those major customers who pay extra to get units with tighter/guaranteed specifications (and a customized label). Those units are selected from the general pool. The open market gets those which don't quite meet those tighter limits, but also - and that might be the overwhelming majority - all those units which haven't been tested at all. You're free to purchase a set and select the best and sell the rest on Fleebay (which I think some do).
Does an electrolytic (or tantalum) capacitor need a minimum DC bias in order to work properly (EDIT: to be properly "polarized")? I ask while the top capacitor in the bootstrapped combo sees "0V" best case.
The drift rate at that temperature of 95°C is nowhere specified, but it is an Arrhenius calculation, according to e.g. P J Spreadbury: 'The Ultra-Zener.. is it a portable replacement for the Weston cell?' , Meas. Si. Technol. 1 (1990).So this mean one can expect about a 2 times higher drift rate for the LTZ1000 A due to the about 10 K higher set point needed. This could be a bit less if good insulation is used around the non A version.
They demonstrated, that the drift of the LTZ1000 doubles with each 10°C increase of the oven temperature. At 55°C they measure typically -2ppm/year. 95°C would yield a 16 times higher drift.
....
Due to the square law for the heater, half the power needed means about 70% of the current needed. So there is some power saving for the A version possible, but not that much if external insulation is added. A first point to save power and keep the drift low is choosing an oven temperature that is not too high for the planed use.
Just as a crazy idea: The heat lost from the transistor to drive the heater could in theory also be used to heat the reference - so maybe have the transistor on top of the LTZ instead of somewhere far away. This may need a modified compensation however, to really work at the low power end.
I doubt that one can calculate individual noise from pairs.
Indeed you can :) Click :
www.intersil.com/data/an/an177.pdf (http://www.intersil.com/data/an/an177.pdf)
The calculation is not difficult!
The measurement is a pairwise sample DUT noise density over frequency. Because the assumption is that the samples are statistical independend the noise power at sa specific frequency is simply the addition of the noise power of the DUT pair.
If you have n=4 samples you can make n*(n-1)/2=6 pairwise measurements. I will call the reference sample spectral noise power at a specific frequency simply a,b,c,d.
What you measure with the pairs is the sum of the noise powers
a+b
a+c
a+d
b+c
b+d
c+d
You get the noise power of the first sample reference by the following expression:
a=(1/3)*(((a+b)+(a+c)+(a+d))-(1/2)*((b+c)+(b+d)+(c+d)))
a=(1/3)*((3*a+b+c+d)-(1/2)*(2*b+2*c+2*d))
a=(1/3)*((3*a+b+c+d)-(b+c+d))
a=(1/3)*(3*a)
a=a
For the general case n the expression is:
a=1/(n-1)*(sumwith(a)-1/(n-2)*sumnot(a))
The application note from LT does not give a calculation example. It is only stated that you have to use a much better reference as a DUT partner.
Calculating the noise of individual reference from measurements in pairs is rather difficult, as the noise is not constant in time :-DD. So the individual noise amplitudes will scatter quite a bit. So the cornered hat version is not really practical, as the individual measurement will contain quite scattering.I seem to remember seeing this somewhere on here where someone had 5 LTZ1000 references that were averaged (must be nice). Is there a multiple LTZ1000 (or even two) reference design pcb on eevblog?
There are usually 2 practical method to check the noise:
One is to amplify the noise with an AC coupled amplifier. So there will be lower frequency cut off, at maybe some 0.1 Hz.
The other method is using a known lower noise reference to subtract. A first approximation could be using the average of several (e.g. 5) references as the point to compare too. Once really bad units are out, one can assume the average is lower than the individual units.
R5, 200K looks like too high a value. the LM399 operating current is 0.5-10 mA, so with a 24V supply, R5 should be between 34K and 1.73K.
The schematic does not show the connection between Z- and GND.
R5, 200K looks like too high a value. the LM399 operating current is 0.5-10 mA, so with a 24V supply, R5 should be between 34K and 1.73K.Here you go; and an update on the PCB layout
The schematic does not show the connection between Z- and GND.
but I decided against it this time, opting in for through hole parts
Quotebut I decided against it this time, opting in for through hole parts
Go for TDP1603 10k network then or use a NOMCA1603 network the same way and use a SMD adapter board, to make it throughhole. Still much cheaper.
10k:25k gives you the right 0.4 ratio you want and all resistors within the package can be used to form that ratio.
-branadic-
TCR: Tracking ± 5 ppm/°C (typical) -55 °C to +125 °C
Stability: RatioΔR ± 0.015 %1000 h at +125 °C
Shelf Life Stability: RatioΔR ± 0.002 %1 year at +25 °C
Looking at the data sheet these nomca networks may not be good enough?
I ordered the PCB with oshpark, I will share the project and gerbers after I tested it.That is correct. The actual zener current is set inside the LM399 and due to the regulation loop inside the output impedance is quite low. So the current setting resistor is less important (some factor of 10, maybe more) than with a zener direct.
Just to validate my understanding of the resistors in the LM399 circuit, I understand the ratio setting ones very well. My understanding is that the stability of the 200K resistor is irrelevant, and that the 5K current setting resistor has very little impact. Based on the LM399 resistor thread. Is this correct?
Price is of no issue, that's why I was leaning towards VHD200, I'm not planning on selling any of these, they are for me, and me aloneQuoteLooking at the data sheet these nomca networks may not be good enough?
Please refer to the thread Statistical arrays (https://www.eevblog.com/forum/metrology/statistical-arrays/msg2026729/#msg2026729), they behave much better than the datasheets indicates.
There is a lot of knowledge on this board, that is worth checking.
-branadic-
Hello John,
I'm considering placing the LM399's upside down
but no reason not to make it as noise free as practical.
You may try to simulate the dynamic response of the 399 to the output capacitance. Below a model I put in LTspice (the TC characteristic is an example only as you need to know the actual parameters of the zener and other components on the chip, the schematics comes from the datasheet).
.. Interesting!! Here's link to some earlier LM399 LTSpice simulations we did back in 5/1/2020 to get a "feel" for how the LM399 behaved.FYI - here are the LM399 and MAC199 die shots with an analysis what is there on the chip. A good simulation model would be nice to have! :)
.. Interesting!! Here's link to some earlier LM399 LTSpice simulations we did back in 5/1/2020 to get a "feel" for how the LM399 behaved.FYI - here are the LM399 and MAC199 die shots with an analysis what is there on the chip. A good simulation model would be nice to have! :)
https://www.richis-lab.de/REF02.htm (https://www.richis-lab.de/REF02.htm)
https://www.richis-lab.de/REF02a.htm (https://www.richis-lab.de/REF02a.htm)
Absolute measurements do NOT exist.Number of people inside a building.
This is the first version schematic.What program allows you to draw such beautiful circuit?
(http://www.bramcam.nl/NA/Cap-DA-Leak-Tester/Cap-DA-Leak-Tester-01.png)
There are 2 common mode chokes (small FT toroids) now.Mhm,
PS: Version 3 - Added 78(L)15 for analog only..
What about R1, R2, R3, R4 values??
The 100nF ser 10ohm - EricM indicated the new ADR1399 is more sensitive to capacitive loads than the LM399..Where can I find the ADR1399 datasheet?
Thanks for sharing. I have another question
4 x the zener current (so likely more like 2 mA to run instead of some 1 mA as the usual LM399 current)
1/2 the noise
lower differential output impedance (so less critical resistor for the current, is already not that critical)
Afaik the zener current in LM399 is set to 250uA.
Current through D3 is held constant at 250 μA by a 2k resistor across the emitter base of Q13 while the emitter-base voltage of Q13 nominally temperature compensates the reference voltage.
..You can even use the substrate diode as a "heater", imho. Say 20mA*0.65V=130mW. TO18 thermal resistance is something like 300C/Watt, 0.13*300 = 39C, with Tamb=25 you'll get Tpackage=25+39=64C.
If it is about a more pulsed operation (e.g. 99 ms zener and 1 ms temperature measurement) the more suitable way would be a switch to turn off the connection when in temperature mode.
[..]Is DA really an issue, if one uses two capacitors in series and charges the mid point from the output of the zener using a 2nd resistor (as described in AoE3)?
For filtering I would not consider electrolytic caps practical, because of the temperature effect and leakage that can change with time and temperature. The DA will also requite extremely long settling - maybe not as long as the LM399, but it would start all over after power off.
[..]
That little trick is designed to deal with leakage currents, not dielectric adsorption.Well, yes, but what is the ill-effect of DA, if not voltage drop over the resistor due to the charging current?
Where can I find the ADR1399 datasheet?
I also saw the "Request samples" button was available, though didn't try it.
I also saw the "Request samples" button was available, though didn't try it.
I works ;D. Delivery date pending...
I collected several MAC199 (LM399 clone in TO18 case) with broken heater. I've tried to utilize the parasitic diode for chip temperature measurement.Bringing the substrate above circuit ground may forward bias parasitic diodes from the substrate to circuit elements like NPN collectors.
Basically it works, with say 80uA fw current the voltage at the heater's ground changes like (aprox)
31C (fingers) 620mV
26C (amb) 637mV
IPA large drop 677mV
There is an issue with it, however - while the 80uA current is injected into the heater's gnd (parasitic diode anode) the Vref (standard 10V Vref like the above) increases by +3000ppm. That is way more than a drop on the bond wire - thus it messes up something on the chip, it seems..
in the ratio 3/7Hmm,
thanks alot for the input andreas
i am unsure how you make the 3/7 or 1.5/3.5 ratio with 8 equal value resistors ?
please explain ?
The weak point of the LM399 is more the popcorn type noise,
6500 is noisy toy, but very stable.Hello,
34410A is very low noise equipment... but poor stability.
Some time ago Ill try to explain about "time delay between points" to Mr.Kleinstein, but my English skills is poor, my explain looks bad.
10.6V is fine as a voltage reference, as is 6.9V. The reason to chose a 'standard' voltage is so that it can be compared directly (difference method) with other 'standard' voltage sources (using e.g. nanovoltmeters with high resolution, but a limited range or a null-voltmeter with an even smaller range).how can this deliver 10.000V out, when the LM399 is about 7V ?You can see exact value on legend into post #1227
No one could prove to me "REF should be 10.00000000000V exactly", therefore, I use a voltage around 10.6V.
10.6V Its fine value, because most DMM accept overload to 10% on 10V range, without any problems. ;)
Probably the main problem is that I'm not a voltnuts, so 10.6V works fine for me. >:D I can calculate ppm drift with 10.6 base w/o problem.
Moreover: Into my last toy, i use direct connection LM399 to 20 bit DAC with OPA189 buffer and mirror... so... as result DAC output is +6.9V...-6.9V, no problem here... works very precise! ;)
TL;DR . Bob pease or jim williams had a circuit where they used like 10 of the lm199's. anyone know where to find that schematic ?
In this thread (above) i reported a build of a 5x LM399 array, with image and schematic including a PWM gain stage 7 => 10 V. There were some measurements on initial aging and a six months log.why pwm ? use a multiplying DAC ... send output of the dac into summing amplifier.
Meanwhile i built a 10x LM399 reference with 14 V output (2 by 5) with a PWM divider 14 V => 10 V and programmable for use as calibrator. Hope to report measurements later.
Regards, Dieter
DACs are not that stable on the long run. Many internaly use resistors for the R2R chain and are thus not more stable than the resistors.
The PWM circuit can be built very stable. This was the result of Fluke research many years ago.
To be honest i looked into this after reading the 2019 reports of Andreas and branadic on their experiments with PWM gain stages (see above). Meanwhile they seem to have better results, too. As far as i remember they initiated kind of a calibration circle in May, using a PWM.
Well, I could guess a HF noize feedthru via parasistic capacitance, but resistor behaves same, and this could be reduced with bead.Hello,
Nothing wrong in using JFET for startup, just additional cost.Well, I had feeling that answer is likely as you described: it definetly makes sense with jfet based reference, as it have pretty high impedance (typ few kOhms), and just useless additional cost for ordinary zener.
I used single JFETs as 3-5V voltage regulators.What a coincidence, my scaler use jfet as zero-current regulator for pwm microcontroller. Of course here and there capacitors on power lines. Just checked noise with ACV mode in DVM, seems all ok.
With enough filtering one could use a boost converter, but it still need effort and care with the layout. It can be done, but the return is not that great. One can as well start with a higher voltage from the transformer / battery. Ideally this would be in a way (e.g. with diodes) to allow seamless change from one to the other. The power consumption of the reference is not very low ( e.g. 200 mW range) and one would thus anyway want a relatively large battery.
If a temperature variation of 29 - 27.9 = 1.1 K gives 57 - 37 = 20 uv this is about 3 ppm/K of 7V after the amplifier, more than expected for a LM399. For the 10 V output the TC is more like 6 ppm/K. Maybe the meter used to determine these graphs exhibits some TC to start with.
The difference between 3 and 6 ppm/K is certainly caused by the voltage divider. What is the TC spec of those resistors?
Regards, Dieter
..and what is your voltage regulator there? The resistors values around it are quite large..
In general 0.1 % resistors are specified with TC of 10 or 15 ppm/K, e.g. PTF56 resistors. So those 1 % resistors you got may be more like 50 or 100 ppm/K and even if they are from the same lot and you use kind of averaging, 3 ppm/K as a residual TC is better than expected. Yes, one can use copper wire to tune the TC. In order to do that one needs some kind of temperature chamber in order to vary the temperature of the circuit while the meter remains at near constant ambient temperature.If a temperature variation of 29 - 27.9 = 1.1 K gives 57 - 37 = 20 uv this is about 3 ppm/K of 7V after the amplifier, more than expected for a LM399. For the 10 V output the TC is more like 6 ppm/K. Maybe the meter used to determine these graphs exhibits some TC to start with.
The difference between 3 and 6 ppm/K is certainly caused by the voltage divider. What is the TC spec of those resistors?
Regards, Dieter
1/2W 207. As per seller specs is about +-10ppm/C. https://pt.aliexpress.com/item/1005006320806775.html
So you mean the circuit is performing as per component specs, and I need to look for strategy to reduce the circuit TC like adding copper resistance on the lower side of the feedback network (and other resistor on the upper side to maintain the 10v)?
..and what is your voltage regulator there? The resistors values around it are quite large..
It is a HT7550. I keept the resistors high to keep low bias current. The circuit is slow due to the time to charge the capacitors. Takes around 1 minute to reach final voltage of 14.5V. I am feeding it with a 5S battery.
The first point to improve would be to improve the ground routing: take out the shart GND wire near the LT1001 and give the heater current a separate return path to the regulator or power connector.
The Vreg variant.. The two 10u input/output capacitors C1 and C2 around the Vreg should be wired to ground. In your wiring the two important decoupling capacitors C1 and C2 are "missing", because they are wired to ground via 300k..
Went back to #1251 to iMo simulations. :/ I guess will use only 22nF on the one misbehaving or even test a different SMD part of 10nF not to make the circuit more susceptible to the supply noise.
..Will update the C1 position and reduce the voltage divider resistances there to make the response fasterYour board will take perhaps 80-100mA when cold, after couple of seconds when already hot the current drops down to perhaps 30-40mA. If I were you I would go with R1=1k and R2=2k (in my Vreg schematics) or something like that.
Do not use more than 16-17V input. It could be your 5S battery is too much (21V max), try with 4S instead (16.8V max).
I remembered Dave video that opamp offset changes with supply voltage. I have two in series. Now I realize why changing from 15V towards 12V the 10V output changes. :) I imagine that same happens in a smaller scale as the battery empties or you have changing load but the LDO keeps this in check for some degree.
I can see "100mV" dropout in my DS (at 1mA out), with 30-40mA it could be more..
Dropout voltage is defined as the input voltage minus the output voltage that produces a 2% change in the output voltage from the value at VIN= VOUT+2V with a fixed load.
1/2W 207. As per seller specs is about +-10ppm/C. https://pt.aliexpress.com/item/1005006320806775.html (https://pt.aliexpress.com/item/1005006320806775.html)
So you mean the circuit is performing as per component specs, and I need to look for strategy to reduce the circuit TC like adding copper resistance on the lower side of the feedback network (and other resistor on the upper side to maintain the 10v)?
People sometimes use the opamp+LT1010 as well..
What is wrong w/ it? LT1010 is 20MHz BW, 75V/us, +/-150mA.People sometimes use the opamp+LT1010 as well..I am not sure what that was a response to.
What is wrong w/ it? LT1010 is 20MHz BW, 75V/us, +/-150mA.
Within the loop with the opamp ("opamp+LT1010") it could provide source or sink capability easily.
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My point was that emitter follower promoted many times in this thread makes more harm than good..
Output impedance of 10..100K is no go for...
My point was that emitter follower promoted many times in this thread makes more harm than good..
Could you elaborate a little bit what kind of harm it may cause?
If somebody knows how the DMM pulses look like we may make the simSee attached screenshot from Keysight 3446x data-shit. "Noise and injected current: Keysight Truevolt DMMs contribute less than 30% of the injected current than alternatives. Compared to some lower cost alternatives, Truevolt DMMs offer almost 100% less noise"
In the same sim schematics and the same pulses: with the transistors the voltage droop is say 12ppm, when transistors are replaced by a faster opamp like the AD8065 there are just 0.1ppm transients left.
In the same sim schematics and the same pulses: with the transistors the voltage droop is say 12ppm, when transistors are replaced by a faster opamp like the AD8065 there are just 0.1ppm transients left.
An class-ab stage doubles the transconductance so should have half of the voltage droop.
I suspect this explains why Sziklai pairs are sometimes used at low current, especially when the first transistor is an FET.
- It must work in forward and reverse polarity on battery as well as transformer powered.
- I want to use the voltage reference in real modern world. I dont blame for fout the DMMs auto-zero implementation or the modern zero-drift OpAmps. I assuem that they exist.
- cope with EMI challenges from modern world - from solar Inverters, to EV chargers. from LED bulbs to BLDC air-conditioning /refrigerator.
Poor quality resistors not only have a problem with the TC but also with long term drift and noise. For a reference source the long term drift is the more serious issue. Even with high grade parts it is hard to get specs on the long term drift and already the possibilty for drift is an issue, as one would not easily know if they drift.
The TC part is only the point that is easy to measure and compare. A resistor array would really be the more sensible choice.
PS: would it be better to thermocouple to the board the resistor network on on the contrary to try to thermally isolate it like with some plastic spacer? :)
The settling time sim - after the same NEG pulse as above in my ADR1001#1 buffer.Hi Imo,
With the POS pulse it is 46us. The CLC filter adds the damped resonance, of course. Not counting capacitance and inductance of the leads.
Without knowing how the typical pulses look like it would be just wild guessing, imho (after 11 years of this thread) :D
..Common mode chokes are not ideal inductance. their core is optimazed for max loss instead of efficiency. It is nice to look at IR images of such chokes.
Common mode chokes also behaves as diferential.
Common mode has parasitic capacitance. Capacitance depend on inductance - number of turns.
Common mode chokes are in many flavors. Depends on application that you want to use it..
The OPA2209 is a rather expensive choice for the output buffers inside the loop. A NE5532 should be about as good though maybe with a little more power consumption.Many thanks for quick and relevant review Kleinstein.
The filters before the output buffers may effect the loop stability. One may want an extra capacitor in direct feedback to get the fast feedback part for U3B and U1B, at least as an option in case it oscillates.
U4 seems to have no power supply.Thanks you save two patches in PCB
I prefer the PTVS11VP1UP,115 Diode for my references up to 10V.I will order a bunch of TVS and select the one with lower leakage across T range
Is there any reason to put R13, R16, R17 outside the feedback loop (R14, R22).I have doubt too. I still did not make a choise between OpAmp stability vs Performance/LowZ
How do you dimension the output capacitors? (for which frequency coupled into the output)Still work ToDo - I put some random numbers :-) Thanks for the tip I will on LT AN86
C4 with 100 pF is rather low compared to ~1 nF of the transient zener.
I would wire C8, C10 not to GND but to the negative Input (PIN 6) of U1B / U3BI dont like placing this capacitors to GND too due to worseing of stability but I was also aware of choper injection back input. I will place them both and decide which to use later on
The LM399 reference is still relatively noisy (e.g. 100 nV/sqrt(Hz) of white noise part and quite some 1/f noise on top) . So the op-amps for the buffer should not be that critical. The question is more if one needs AZ op-amps with there possible spikes or could use a classic precision OP-amp like OP07, OP177 or newer OPA207, OPA205, ADA4077.OPA2186 took my atention due to extremely low price. I'm reluctant to use it blindly since it seems different from other TI modern Az opamps. The max voltage is 24V. All other TI AZ OpAmp members have 36V. Is it different architecture or different Fab process? - I will order few and evaluate them. I like OPA207, OPA205, ADA4077 and especially OPA2205 but I don't like their price.
The LM399 reference is still relatively noisy (e.g. 100 nV/sqrt(Hz) of white noise part and quite some 1/f noise on top) . So the op-amps for the buffer should not be that critical. The question is more if one needs AZ op-amps with there possible spikes or could use a classic precision OP-amp like OP07, OP177 or newer OPA207, OPA205, ADA4077.OPA2186 took my atention due to extremely low price. I'm reluctant to use it blindly since it seems different from other TI modern Az opamps. The max voltage is 24V. All other TI AZ OpAmp members have 36V. Is it different architecture or different Fab process? - I will order few and evaluate them. I like OPA207, OPA205, ADA4077 and especially OPA2205 but I don't like their price.
OPA2140 is another chiose - since one opamp can serve as both - High accuracy, low Ib, and high BW - no need of dual stage opamps. So one expensive instead of two cheap
If you look for a cheap precision OP-amp, the OPA202 / OPA2202 are good candidates. They can kind of replace the OP07 with less power consumption and slightly better performance in most aspects.
I'd guess the 2N2222 with its 800 mA current spec is more robust than most opamps
What are the practical advantages (no theory please!) of having these discrete transistor output buffers over using a buffer opamp?
Just wondering if I should change my design...
What are the practical advantages (no theory please!) of having these discrete transistor output buffers over using a buffer opamp?
It's sad to hear about the MAB399 heater problems. I have 2 pieces, both have been aged for ~1000 hours on a current limited power supply. It was set for about 100mA and that might have protected them on each start. Inrush limiting as suggested by iMo is a must then. Would be interesting to see what failed inside of a reference with dead heater. I don't think this part saw much use because it arrived too late (1990 judging by the date code), when the soviet economy was already down the drain. No use means not much testing, so it could be flawed :(.
..It was set for about 100mA and that might have protected them on each start.I had none current limit and lost perhaps 5 pieces out of a dozen.
..I don't think this part saw much use because it arrived too late (1990 judging by the date code), when the soviet economy was already down the drain. No use means not much testing, so it could be flawed :( ..
P.S. as a kid I had TESLA bass guitar amplifier. It had just average sound, but was built like tank.. Undestructable ... ^-^
PS: My bet all MAB/MACx99 refs available today at ebay or in various eshops were already carefully "sift through" (mind they are 30+ years out of production), also the markings on their hats might be replaced/forged. All chips are "MAC199" afaik, and the "MAB/MAE" versions were just stamped onto the easily replaceable plastic hat after the binning during the testing..
OPA2186 took my atention due to extremely low price. I'm reluctant to use it blindly since it seems different from other TI modern Az opamps. The max voltage is 24V. All other TI AZ OpAmp members have 36V. Is it different architecture or different Fab process? - I will order few and evaluate them.
I am in the process of updating my preferred operational amplifiers list and I have used the low power OPA187 recently.
Thanks, i was looking for a zero drift opamp with input rail to rail and couldn't find one. All i found excluded the upper 1.5 to 3 V of common mode voltage or limited operation voltage to 5 V.
The OPA186 datasheet figure 6.7 seems to indicate some anomaly at the lower 3 V of the input range. Maybe that disappears at lower supply voltage. Anyway now there is a choice..
The f = 0.1 Hz to 10 Hz input noise spec 125 nVRMS for 90 uA supply current is amazing.
I would like to hear more about your list when you are ready. I would suggest adding LTC2057 to the evaluation list - I recently did some evaluation on a group of zero drift opamps, measuring the input bias current over CM voltage range, whilst varying the R/C of input/feedback. LTC2057 was the only device to have both: 1. Nearly constant input bias over CM voltage range, 2. Maintain that constant input bias over a variety of R/C conditions. It was quite astonishing to see how badly behaved other parts were under various conditions and LTC2057 would always have a flat clean line on the charts.
..What is the difference between using a PI bifilar common mode filter on the output vs. using separate cores for each wire like in older voltage standards..
3 separate coils are different. They could have the same effect on the common mode signal, but the effect on a differential signal is quite different, with much more inductance seen to the differential signal...What is the difference between using a PI bifilar common mode filter on the output vs. using separate cores for each wire like in older voltage standards..
Provided all three coils are perfectly identical (incl the winding direction) there is none difference.
Thanks for the suggestions!
I've revised the schematic like so, and added a jumper for a small NTC temperature sensor if needed.
In addition, I added a low pass filter on the output of the LM399 (Is this worthwhile?), and also removed the trimmer stuff. (One lm399 is like ~7.001V which makes it close enough to 7V that I don't need to do much trimming)
The original idea was to ovenize just the resistors (and possibly the op-amp) with a PTC heater, then encapsulate everything in some insulative foam.
The LM399 will be placed separately.
Unfortunately I do not have another 0.1% ~3k resistor to spare, so I'll use 3 0.1% 1k instead. (Originally I wanted to parallelize 3 10k, but I don't have that)
For the buffer, I think it doesn't need to be high current? I only intend for it to be a voltage reference for testing purposes.
As for the rectifier, I've decided to use a discrete module, which seems to take AC 18V and turns it into DC 24V. I may possibly need to find a small enough transformer to put into a case.
A few questions:
How can I know the current flowing through the zener diode based on the resistance?
Also, why should U4 be located near the PTC?
For the buffer, I think it doesn't need to be high current? I only intend for it to be a voltage reference for testing purposes.
..
Watch out that LM399's 'nominal' current source is 1-2mA in most designs (although the datasheet does indeed note up to 10mA) whereas the ADR1399 datasheet has bumped the recommended current up to 3mA (up to 10). In truth, for a design trying to accomodate either, I'd guess either of them would run fine at 3mA.
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