PROJECT: Micro-Voltmeter Design

[RandallMcRee]

Yes, this is very common. Most (non-audiophile!) capacitors have steel leads, tinned. I don't know but I think it has something to do with the manufacturing process requiring a weld of some sort.

I always check just as you did, with a magnet. Also check critical resistors--you never know!

In most metrological circuits I don't think its a big deal (that is to say at DC), but something to be aware of.

[Kleinstein]

With wires from 100 % tin one would hardly be able to solder with lead free solder. So it is 100% surface only. For normal use the capacitor wires would not matter. Just avoid using the wires also in the DC signal path.

[Lesolee]

I always check just as you did, with a magnet. Also check critical resistors--you never know!

Sadly I do know. Every ordinary metal film (and even old carbon film resistors) I have, have magnetic end caps. 

1N4148 glass bodied diodes, magnetic. (Maybe Kovar due to the glass seal.)

So I have no ideal solution for my front end resistors since they have magnetic end caps. Nichrome is supposed to be non-magnetic, but it is not well matched to copper in terms of Seebeck coefficient. It is not at all clear if there is such a think as a low thermal emf resistor (although low resistance current shunts do mention it.) The datasheets just say "end caps" without mentioning their material. Having bunged out big bucks for the low thermal EMF binding posts, it is annoying that the resistors now have this built-in error mechanism. 

[Kleinstein]

Some NiCr alloys (with some additions) are additionally optimized for low thermal EMF.  For precision circuits one should have a symmetric thermal design for the critical resistors, so that the temperature is about symmetric and thus no net thermal EMF. This could be a bit tricky with a free air / proto-board circuit.  With only µV to aim for this is not so bad. With the ADA4522 one could however increase the gain even further. The noise level is low enough to still resolve some 100 nV and with some filtering / patience even 10 nV.

The overall power of the circuit is low and the power of the 1 critical resistor is also relatively low if the voltage is low. So I would not worry so much about this

[Lesolee]

Here it is, the functioning Mk2 design. It is not fully tested yet as the new 18650 batteries failed to arrive last week. (As mentioned in post #25, the old ones were rubbish. I measured them as 400 mA.h as compared to their stated spec of 4200 mA.h  - marked as UltraFire, but may be fake  ).







Kleinstein’s post (#27) showed that the chopper amp bias currents have a dubious relationship with impedance, which is highly undesirable. And Splin (post #13) suggested considering an ADA4625, which I have used.

Basically the box, terminals, pot, DVM module, and charger socket are all as they were, but the two amplifier boards have been replaced. In the process I planned from the start to use Magic’s suggestion (posts#3, #10) to lift pin 3. This was not mechanically sound on the earlier design. It is fine in this new, more mechanically stable configuration.

There is a major gotcha with the ADA4625. The worst case offset voltage is shown clearly as 80 µV, with 15 µV being more typical. GOTCHA. Turn the page for 5V operation and it suddenly becomes 600 µV.  My nicely laid out front end had to have a hasty offset correction pot bodged in. The uncorrected offset was around 200 µV, so the value of R20 is on the edge of workability (or tuned to perfection if you prefer).

The output of U4 is trimmed to 0 V when the main input terminals are shorted, then the set 2 V pot (R3) is adjusted to make the display read 2.0000 V.

The bias current measured as around 5 pA using the 1M//100nF method. The bias current noise was about the same as measured on the Mk1 design. But then I had another rare moment of clarity. The bandwidth of the amplifier is around 1 Hz. The rms Johnson noise in a 1M resistor in a 1 Hz bandwidth is 0.127 µV, which is around 0.8 µV ptp. That is almost all of the noise I was seeing on the 1M//100nF test.

I checked the Fluke Calibration: Philosophy In Practice manual and was disappointed to find that whilst “bias current” is indexed, they were short of an actual procedure to measure it.

I used the “charge a capacity” method, which I have never previously used, and have no idea where I heard of it. 

 Mk2 bias current.xls (18.5 kB - downloaded 45 times.)

The charge slope of 2.5 µV/s into a 2.2 µF capacitor gives a bias current of 5.5 pA, in good agreement with the 1M//100nF method. 

[EDIT: updated circuit to v1.10] March 25, 2020
[EDIT: updated circuit to v1.30]

[David Hess]

Diode conductance at zero volts is 20 to 30 mS/mA * leakage which for a 1N4148 is consistent with your 300 Mohm measurement.  A true low leakage diode would be more like 300 Gohms.

It would be instructive for you to explain where this value has come from.

Certainly I would agree that a well biased base-emitter junction would give 26 ohm at 1 mA as an equivalent output resistance (to which a bulk resistance of say 3 ohms could be added). That would equate to 0.04 S/mA. But I would not be so bold as to expect that to work correctly at or near zero bias.

Bob Pease discusses it on page 69 of his book Troubleshooting Analog Circuits.  I measured values consistent with it once and so is dietert1's result.  I do not know the details about how Pease arrived at that value so I will quote what he wrote below:

Of course, you don’t have to reverse-bias a diode a lot to get a leakage problem.
One time I was designing a hybrid op amp, and I specified  that the diodes be con-
nected in the normal parallel-opposing connection across the input of the second
stage to avoid severe overdrive (Figure 6.3).  I thought nothing more of these diodes
until we had the circuit running - the op amp’s voltage gain was falling badly at 125
“C.  Why? Because the diodes were 1N914s, and their leakage currents were increas-
ing from 10 nA at room temperature  to about 8 uA at the high temperature. And
remember  that the conductance of a diode at zero voltage is approximately (20 to 30
mS/mA) X  ILEAKAGE.  That means each of the two diodes really measured only 6 kohms.

Because the impedance at each input was only 6 kohms,  the op amp’s gain fell by a
factor of four,  even though the diodes may have only been forward or reverse biassed
by a millivolt. When we substituted collector-base junctions of transistors  for  the
diodes,  the gain went back up where  it belonged.

Thus you cannot safely  assume that the impedance of a diode at zero bias is high if
the  junction’s saturation current is large. For example, at 25C  a typical IN914 will
leak 200  to 400 pA even with only 1 mV across  it. Therefore, a 1N914  can prove
unsuitable as a clamp  or protection diode - even  at room temperature - despite
having virtually no voltage biassed across it,  in even simple applications  such as a
clamp across  the inputs  of a FET-input op amp.

[Lesolee]

I thought I would practice my diode leakage current measurement technique on a rubbish 1N4148 diode, just to get warmed up, so to speak.

Here is my measurement circuit:



And here is the setup, with particular emphasis on the cardboard box, cut out carefully to exclude both air flow and light. This is particularly important on a 1N4148 which has a transparent glass body (and is therefore totally unsuited to low-leakage applications!).



The measurement noise has been greatly reduced since the last tests simply by putting 2µ2 across the 1M resistor (rather than the 100 nF used previously). This kills the Johnson noise nicely.

The point of interest is that regardless of forward or reverse bias, at these levels (up to ±2 mV) it makes little difference if the diode is forward or reverse biased.



It should be noted that the 31 pA/mV slope (= 32 Meg) is in poor agreement with dietert1’s mention of 300 Meg for a 1N4148 (post #11).

It should be noted that the leakage current is proportional to the applied voltage in this very limited range around zero, but this leakage resistance is strongly dependant on the temperature.

Now that I have had a little practice, I can actually try some proper low leakage diodes (and diode replacements). 

[EDIT: major scaling error in mV applied.   Was a factor of 10 too high.]

[EDIT: Just to be clear, nobody in this thread has ever suggested that using a 1N4148 as a clamp diode is a good idea. The mentions of the impedance of the 1N4148 were not presented as recommendations.]

[EDIT: Post #66 below clarifies Bob Pease's formula such that the ILEAKAGE is the reverse saturation current of the diode. Remove comment about the formula being wrong, and add some detail. ]

[Lesolee]

I changed the top resistor in the bias divider chain down from 1M to 10K so I could extend the bias range, in order to possibly see something interesting.



Clearly, at current levels below 1 pA the readings are uncertian. The diode curve agreement on the log plot is very nice.



Probably the other good options for diodes will be difficult to compare since they will all probably be excellent at room temperature. It may be necessary to heat them up to separate them out in terms of the best leakage performance, although the BC550C is demonstrably excellent for this application.

[EDIT: updated plots to include an 82°C measurement as well, and to correct the transistor type to BC550C (not B)

[dietert1]

Apparently you did not perceive the recommendation Kleinstein and i made to use BAV199 low leakage diodes with a small negative bias. No reason to start false communication.
Maybe you can have a look at the Fluke 845A input circuit. They implemented a pair of antiparallel diodes similar to your original proposal. But they bias those diodes from a copy of the input voltage that they obtain from the output of the chopper amplifier output using a voltage divider. Yet another way to implement offset currents below 0.1 pA. You can find the schematic in one of the Fluke 845A threads.

By the way, about nine months ago i used a 100x ADA4522 preamplifier with one of our HP 3456A voltmeters to compare voltage references. Allan variance with shorted input showed a broad minimum of 3 nV at about 30 to 60 seconds.

Regards, Dieter

PS: The advantage of a BAV199 over a pair of separate diodes is thermal coupling to keep leakage in balance.

[Rerouter]

Lesolee would you be able to try the base-collector as a diode, seems that should be the lowest leakage configuration, and curious about the forward voltage hit.

[Lesolee]

Lesolee would you be able to try the base-collector as a diode, seems that should be the lowest leakage configuration, and curious about the forward voltage hit.

Yes, MadTux suggested a C-B junction with B-E connected together (post #28).

Post #55 (Bob Pease via David Hess) suggests a C-B junction with no mention of the emitter leg.

I am inclined to cut the emitter leg off.

But the bigger issue is discriminating between the different diodes and/or structures. Below 1 pA is largely irrelevant to me in this application, as is high temperature operation. Nevertheless, it is interesting from an academic point of view (and for other applications). So I am just trying to think how to improve the discrimination, perhaps by heating the "diode" up to say 60°C. I just need a consistent way to do so (that does not involve a mineral oil bath).

[Rerouter]

For me the leakage is already brilliant, its more what the forward changes is as that should be a larger junction, (thus lower leakage)

The emitter is used when you want a slightly faster reverse recovery time, about 200ns vs 30ns from other tests I can find. I would assume this would also make the leakage a bit higher.

This should also imply the C-B junction leakage should decrease for higher voltage rated transistors assuming similar beta.

[Lesolee]

The emitter is used when you want a slightly faster reverse recovery time, about 200ns vs 30ns from other tests I can find.

Where are these other tests? 

I went for a high Hfe transistor on the basis that if it has high current gain, less base current is wasted in shunt resistance. I also tried a 2N3703 earlier (with the noisy test method and Mk1 design) but the test method was not adequately discerning.

A new thermal jig is in progress using bits and pieces on hand. The main body is an earth pin from a non-rewireable BS1363 plug I found in a scrap bin! 

Needless to say I first had to wire the µVM up to an old plug-top power supply to free up a bench power supply, since I only have two single-output PSUs.

[EDIT: found this whilst waiting for the epoxy to set]


from OP77 datasheet as mentioned in AN018 low-leakage diode note from Rod Elliot
https://sound-au.com/appnotes/an018.htm

[Rerouter]

https://sound-au.com/appnotes/an018.htm
https://electronics.stackexchange.com/questions/259713/can-i-abuse-a-transistor-as-an-esd-protection-diode

The second one is more where my guesses come from as it seems to increase capacitance when the emitter is connected, though I will agree it does not show the exact configuration.

[Lesolee]

I found the same app note. Sadly (for you) I posted the link 5 minutes before you. 

This is the donor UK mains plug with the WIP thermal stabiliser. The extra hole was drilled and tapped M3, and the intention is to recycle the fuse holder into a clamp.



And here is the finished article:



That is a 10R 5W resistor, and it was run at around 4 V to get the measured block temperature up to 82°C. The graphs in the earlier post have been updated to include the 82°C data.

[Lesolee]

Lesolee would you be able to try the base-collector as a diode, seems that should be the lowest leakage configuration, and curious about the forward voltage hit.

Here it is, head to head using the same transistor. The temperature was measured on the block, so the junction temperature may be a little less due to cooling through the leads. But both configurations have the same transistor, and the mounting method was not changed for the test, which is the best one could hope for.

I am surprised by the result. The graphs show no appreciable difference between the two configurations in terms of currents.

[David Hess]

It should also be mentioned that Bob Pease’s formula (see post #55) is remarkably inapplicable given that the leakage resistance is independent of current in this range. 

I think some context is missing from what I quoted and Pease is referring to the reverse saturation current of the junction and not the leakage at any particular voltage.  Obviously the leakage at exactly zero volts is zero.

[Lesolee]

The Pease formula (post #55) is easy to derive, now we know what it is trying to say.

[Lesolee]

It is perhaps worth a little look back at a proposed clamp solution using BAV199 diodes. I originally had such a scheme in mind (but not with BAV199s) but rejected it as it was unnecessarily complicated. I also rejected the suggestion of BAV199s as they are surface mount parts and would have been a pain to use compared to the simplicity of a pair of back-to-back transistors.

But let’s just take a look at a BAV199 data sheet snip (the full data sheet is attached as well).



At a reverse voltage of 3 V they typically have a reverse leakage of 200 pA. Doesn’t sound very good to me. 

Oh, maybe I used the wrong manufacturer. How about this one?



That’s better: 3 pA at room temperature, provided you read the “correct” curve. The typical value is 3 pA and the worst case is 5 nA. A factor of 1666 between typical and worst case. That seems like a bit more than 3 sigma! 

Let’s try somebody else.



Wow, 1 pA with a bold line. Why worry about a dashed line. That can’t be important, can it? 

I would love to speak to an in-house engineer at one of these manufacturers as to what on earth we are supposed to do with such a ratio between “typical” and worst case. If I get a reel of all the same performance devices at 1 nA leakage, all my production run of discombobulators are going to be rejected and end up on my bench to fix. But the manufacturers can insist that the device is in-spec. Everyone else got the good ones. 

If you want a well specified diode for low leakage we have the FJH1100. 3 pA at room temperature, guaranteed. WOW! Of course if you use it at any temperature other than 25°C there is no spec at all.  No curves, no typicals. And it costs an arm and a leg to buy. 

Another point, if you look back to the Diodes Inc curves above, their diodes don’t follow the diode equation. They tweak the doping, and the diffusions, and so forth, to make it a good diode. But that doesn’t make it follow the diode equation!

[David Hess]

I would love to speak to an in-house engineer at one of these manufacturers as to what on earth we are supposed to do with such a ratio between “typical” and worst case. If I get a reel of all the same performance devices at 1 nA leakage, all my production run of discombobulators are going to be rejected and end up on my bench to fix. But the manufacturers can insist that the device is in-spec. Everyone else got the good ones. 

It comes down to the cost of testing.  The parts are only tested to meet the worst case specifications, which in this case is the maximum leakage because testing leakage at a lower level costs more time which is money.  Testing time is literally 10s of cents (as of 1980 anyway) per second.  If you want a good example of this, check the price difference on the LMC6081 ($2.45) and LMC6001 ($12.24); they are the same part tested to different leakage specifications.

If you want a well specified diode for low leakage we have the FJH1100. 3 pA at room temperature, guaranteed. WOW! Of course if you use it at any temperature other than 25°C there is no spec at all.  No curves, no typicals. And it costs an arm and a leg to buy. 

If you want an inexpensive but tested low leakage diode, then the 10 picoamp 2N4117 and the 1 picoamp 2N4117A low leakage JFETs are the best option.  Or grade your own parts, I like 2N3904s, or design your circuit so that it can test itself and the parts can be replaced if necessary.

[Lesolee]

If you want an inexpensive but tested low leakage diode, then the 10 picoamp 2N4117 and the 1 picoamp 2N4117A low leakage JFETs are the best option.

Probably the price went up when you weren't looking.

£11.94 each (from Mouser.co.uk) is not inexpensive 

The leakage is also only specified at room temperature. I think this sort of part is no longer popular, and they are being discontinued, or re-manufactured at a significant premium.

[Kleinstein]

The 2N4117 is the old TO18 case. There are newer SOT23 equivalents - not sure about the test level however.

For a hobby project one can just try the BAV199 - chances are very good they are more close to typical. For a larger series: Check a few units per real, and take the risk for rework, or get some custom tested parts or other tested low leakage diode. They are not cheap though.

[Lesolee]

It comes down to the cost of testing.  The parts are only tested to meet the worst case specifications, which in this case is the maximum leakage because testing leakage at a lower level costs more time which is money.

That might actually be reasonable. They realistically expect the parts to be that good, but they can't test to that level routinely so they won't spec it. In that case you would hope they would test-screen a few diodes from each wafer, or each wafer batch, to ensure they are as expected -- but they don't mention that. It would give (me) more confidence if they did. But it does mean that not all BAV199s are created equal. The Diodes Inc data sheet shows a definitely worse diode.

[dietert1]

10 nA (worst case at 70 V and 70 °C) means less than 1 pA at 1 mV. What's the problem?

Regards, Dieter

[Lesolee]

10 nA (worst case at 70 V and 70 °C) means less than 1 pA at 1 mV.

Well it's interesting that you should say that. I would argue that you are applying an unguaranteed extrapolation method to make such a calculation.

How did you calculate that? (And is it <1 pA at 1 mV and still at 70°C?)