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Help needed for DCM SMPS using uC3842AN

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WyverntekGameRepairs:
Hi there! I'm Sterling Ordes, 18 years old, and I'm trying to start a video game console & peripheral repair / refurbishment / accessory manufacturing business. I'm doing quite well so far, getting ready to purchase a business license and other licenses I will need to get me started, so I'm raising money through small partnerships with other businesses and producing my own console accessories. The current product I am designing is a switch-mode power supply for the Nintendo 64.
I have based my design off of the Playstation 1 SCPH-5501 SMPS, since it is simple and has multiple outputs. It uses the uC3842AN high-performance PWM controller IC as the control IC, and is both easy to understand and high-quality.

However, I am stumped when trying to draw up a schematic, as several parts of the PS1 schematic has wiring choices that are... Questionable, to say the least. I'll list them here:
1) The output ground on the low-voltage side is linked directly to neutral via a 6.8M 1/2W resistor (R120), as well as the high-voltage negative / ground (from the rectifier) via 1500pF 125V capacitor. While the capacitor makes sense, as it is used to shunt the high-frequency interference and such to neutral / ground but still maintain isolation, the resistor really throws me off. It is high resistance, but it is visibly connected from the main ground on the low-voltage side *directly* to the neutral, no capacitor to provide isolation. Wouldn't this break the isolation and render the isolation useless? Or is this actually a great way to help surpress interference? Or is it possibly required for the error amplifier to function on the uC3842AN chip? That design choice throws me off, as I am mainly a hobbyist and haven't really gotten into the true nitty-gritty of engineering until very recently, so I don't understand what effect doing that would do to the overall performance and safety of the circuit.
2) In the PS1 PSU schematic, the COMP is connected to the Vfb on the uC3842AN via the optocoupler, along with the capacitor for filtering (if I'm not mistaken). However, the Vfb is also connected to the negative of the mains post-rectifier smoothing capacitor, serving as the ground for the IC, the capacitor linking Isense to negative, and the RT/CT capacitor to negative (as well as a few other things). The Texas Instruments functional diagram shows COMP connected to Vfb through a capacitor and resistor (which would be replaced by the optocoupler), but they are not connected to the main ground / negative rail. What is the significant reason for linking the Vfb to ground? Wouldn't it be prone to interference from other voltages on the ground rail?
3) On my PS1 PSU schematic diagram, the N-channel MOSFET that switches the primary of the transformer seems to be reversed. The source is connected to negative, but the load is connected to the coil and to the positive rail. The MOSFET looks like it is being negative-biased using a 10K resistor connected between the Source and Gate. The Source and Load of the MOSFET are connected via 100 Ohm 1W resistor in series with a 220pF 1KV capacitor. The Load of the MOSFET is not only connected to the primary coil of the transformer, but is also connected to a RMPG06J (I believe general-purpouse rectification) diode, which is then connected to a 100K 1/2W resistor in parallel with a 0.033uF 400V capacitor... which connects to the positive rail (noting that the primary coil is connected to the positive rail as well). What is going on here?

I may have a few other questions, but I will leave this post at this for now. If you have experience with the IC I am using in this design or have experience in switch mode power supplies, I welcome any suggestions as well. Thanks for reading this long wall of text and (hopefully) helping me out.  :)

MagicSmoker:
A picture says a thousand words - quite literally in this case - so sketch out what you think is the circuit and post it here (taking a picture of a hand-drawn schematic is fine as long as it is legible).

The 6.8M resistor across the capacitor that bridges input and output commons is unusual, but it doesn't necessarily break isolation according to the usual regulatory agencies as they set a limit for how much current from the mains can be injected into (either) ground from any Y-capacitance (ie - capacitors from mains to ground, or ground to ground). The worst case limit of 0.25mA (for medical devices) allows for an effective resistance between grounds as low as 960k (240Vrms / 0.25mA), so 6.8M is technically fine. Why it is there, anyway, I have no idea. I guess it ensures the 1.5nF y-capacitor discharges once power is removed, but this isn't really a shock hazard in the first place.

For everything else a schematic would be very helpful. It is also isn't entirely clear what you are proposing w/r/t the transformer; you almost always have to design the transformer for an offline (ie - mains-supplied) SMPS, and while it can be bewildering the first few times you do it, it's not quite the dark art most people think. Okay, it is very much a dark art, just not Voldemort-level dark...

T3sl4co1l:
1. See above, Magic's covered everything :)

2. If this is the typical circuit you're referring to, then it helps to understand how the controller is being used.  The controller has an error amp inside, set to adjust VFB to 2.50V.  It controls the CMP pin.  The CMP pin in turn is read by the current comparator, this being the setpoint for where to turn off the switch, implementing peak current mode control.

So, in short, V(CMP) ~ Iout.

In a common-ground application, the error amp is used directly, and there is a voltage divider from output to VFB, and a compensation R+C from VFB to CMP.  The compensation accounts for the phase shift of the converter and output filter cap, stabilizing the output.

A note about control: you'll often see descriptions of control systems in terms of "if-then" events, or "greater-lesser-than" digital comparisons.  This is wrong: control is a continuous, linear process.  (Well, it can be done by comparison -- call hysteretic control.  But this case happens to be the linear kind.)  The aim here is to adjust the output, from moment to moment, gently enough that it doesn't run away and oscillate, while -- over the long term -- adjusting it so that the output is as close to the setpoint as possible.

Anyway, that's common ground; what if our output is isolated?  We can use an optoisolator to communicate the output voltage to the controller.  But these are not accurate devices.  We can put a resistor to the LED, driving it with a current proportional to the output voltage, and measure a current at the phototransistor that's -- who knows, maybe 50%, maybe 300%!  It varies with part number, manufacture (gain is not necessarily consistent from part to part) and age (primarily the LED fading with use, over 1000s of hrs).  So that's pretty useless.

So what we do instead, is use the controller as a transconductance amplifier.  That is, it's making a current (our output), and we adjust that current by the input voltage.  So it has gain in units of A/V (conductance), and the amps and volts are measured in different locations, hence "trans".  Transconductance.  We set that voltage with the opto, and set the opto's current* with a separate error amplifier -- this time referenced to the output supply, so we can get an accurate voltage at the output.

*Okay so the opto has current gain, converted to voltage with a resistor, and back to current again by the converter, so this is overall a current amplifier now.  If it matters. :P

The key is, by placing the opto inside the control loop, yes it might be in error, but we are controlling that error, and as long as we've properly placed the min/max limits of the system (opto at minimum CTR (current gain) is still able to deliver full voltage to the controller; opto at maximum CTR does not cause the control loop to go unstable), we're golden -- the error is absorbed by the error amp.

And for that error amp, we have a convenient device, a TL431 usually (and many relatives).  This is drawn as an adjustable zener diode, whatever the hell that's supposed to mean -- but the characteristic is nothing like a zener, in fact it is a three-terminal op-amp with an open-collector output and an inverting input pin with a massive (but conspicuously stable) offset voltage of 2.50V.  Seen in this light, we don't need to have REF tied to "cathode" by a voltage divider (the standard "adjustable zener" configuration), but we can use a whole system to provide that feedback, and just like the error amp on the UC3842, we put an R+C across it ("cathode" to REF) to compensate its response.

So that's the complete system -- we can configure the UC3842 as a simple current amplifier by either strong-arming its CMP pin directly (which can't source/sink much current, so is easy to overdrive; in this case, VFB can be tied off to GND or something), or by wiring it as a unity gain inverting amplifier (say 10k from CMP to VFB, 10k from VFB to opto).


3. Not sure what you mean about reversal or biasing -- it's a class D circuit, switching, no bias needed.  It's designed to spend as little time as possible in the linear range, to save on power dissipation.

The usual configuration is a low-side switch (source and gate reference to GND, give or take), with the primary in the drain circuit (from drain to +V).  The peak voltage (when the MOSFET turns off) shoots above +V, driven by the transformer's inductance.

It sounds like there's an R+C damper (or snubber), which probably absorbs the free ringdown of the transformer's inductance with the MOSFET drain capacitance.  This helps reduce lower frequency EMI, at the expense of some power dissipation (mostly in the resistor).

It sounds like there's also a peak clamp snubber: when the drain voltage shoots up (flyback), the diode clamps that voltage into the 33nF cap (which is referenced to +V, because the primary is).  Actually once the diode is conducting, it's a resonant circuit of Lp (primary inductance) and 33nF, but this is so much lower (in impedance and resonant frequency) that it looks like a flat topped waveform, i.e., it seems to be clamping the voltage peak and that's it.  The 100k then bleeds away the excess charge the capacitor has picked up; this is wasted, costing efficiency (there are configurations where this can be saved -- namely, the two-switch flyback or forward converter; these are more complicated to construct, though).

The clamping wouldn't be necessary if the transformer were ideal, but real transformers have leakage inductance as well.  This appears as inductance in series between primary and secondary.  In effect, the primary can't instantly tell what the secondary is doing, but rather that action is communicated through this inductance.  As a result, when the secondary side diode turns on, most of the transformer inductance (charged as it was by the MOSFET) is clamped into the output filter cap (delivering useful power), but a fraction of that energy is trapped in the leakage inductance, and causes additional peak voltage on the primary side.  This can destroy the transistor, or at least require a higher rated one, so it's better to clamp it instead.  It's also wasted power -- there's no way to divert that leakage energy into the secondary, you're already doing all that you can on the secondary side.

So the best way to save this loss, is to optimize the transformer design so it isn't storing a lot of energy as leakage inductance.


4. Hah, nah, resistor dividers and pass regulators are 100% (or more) waste.  Use multiple outputs -- say, a transformer with a 12V and 4V secondary, and then use an LDO to drop 4V down to 3.3V with good stability.  Logic supplies can be picky, and cross-regulation (that is, the amount one supply's voltage changes, due to a change in load current on the other) tends to be mediocre, so it can be worthwhile overshooting just a bit and still using a lossy regulator but only for a volt or so (so the overall efficiency is still good).

If you can't get custom transformers made this way, you can stack transformers (at some cost to performance).  For example, use "12V" and "3.3V" transformers, wired in parallel at the primary; the total load inductance is the parallel combination, so choose them appropriately for nominal load.  Or, use a single "12V" transformer, then put a trans/autoformer in parallel with that secondary, to divide it down to "3V" or "4V".  You can use one 1:2, or a pair of 1:1, coupled inductors in this way.  There are also multi-winding configurable transformers out there, at some additional cost.

Or, at worst -- just make two independent converters.  Can share the EMI filter and bulk (DC) cap.  Can even buy off-the-shelf modules instead of building your own -- a great way to start if you're unfamiliar with safety regulations!

Cheers,
Tim

WyverntekGameRepairs:

--- Quote from: T3sl4co1l on December 09, 2019, 12:07:02 pm ---1. See above, Magic's covered everything :)...

--- End quote ---

--- Quote from: MagicSmoker on December 09, 2019, 11:03:56 am ---A picture says a thousand words - quite literally in this case - so sketch out what you think is the circuit and post it here (taking a picture of a hand-drawn schematic is fine as long as it is legible)...

--- End quote ---

This is all really helpful, and helps me understand and make sense of things a lot better. Thank you so much! As for Magic, I'll post a link to the manual. The page numbers on the PDF that you need to go to are 51 and 52. http://psxdev.ru/files/manuals/SCPH-5500_5501_5502_5503.pdf
I hope that helps!

Thanks, I really appreciate your help :)

MagicSmoker:

--- Quote from: WyverntekGameRepairs on December 09, 2019, 04:43:51 pm ---This is all really helpful, and helps me understand and make sense of things a lot better. Thank you so much! As for Magic, I'll post a link to the manual. The page numbers on the PDF that you need to go to are 51 and 52. http://psxdev.ru/files/manuals/SCPH-5500_5501_5502_5503.pdf
I hope that helps!

Thanks, I really appreciate your help :)

--- End quote ---

T3sl4co1l already surmised much of the functionality here, so some of this will be redundant.

* The topology is a plain old isolated flyback converter.
* R119 & C112 are an RC snubber (or as I prefer to call it, a damper) whose purpose is to suppress oscillation/ringing as a result of the MOSFET's output capacitance and the transformer's leakage inductance.
* D105, C105 & R105 form a self-tracking voltage clamp. Every time the switch turns off energy stored in the leakage inductance flows through D105 to charge up C105, which R105 slowly bleeds off. It's called self-tracking because the clamp voltage scales with power throughput and is always a certain amount more than the incoming DC bus voltage for a given power (based on the energy balance between LI2 and CV2).
* IC201 is better known as the (somewhat infamous) TL431 adjustable shunt reference and it is being used as a combination 2.5V voltage reference and operational amplifier to control the current through the LED in optoisolator PC101. Many consider this a hack (including me), but it is undoubtedly the most common way to regulate output voltage in cheap commercial switchers.
* Q202 injects a little bit of current from the 8V rail into the feedback circuit to improve cross-regulation with the main 3.5V output. This trick is usually applied to forward converters because they have much worse cross-regulation of multiple outputs compared to flybacks; I wouldn't bother with it here.
* R106 is for sensing primary current while R109/R110 and C107 form a light filter to round off any leading edge spikes that might cause the current-mode controller, IC101, to terminate the switch on-time too early.
* IC202, marked as RST598INR, might be a typo for PST598INR. Either way, this is clearly a supervisor IC that keeps a processor in reset until the power supply voltage(s) is correct.

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