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How to calculate UC3842 components?
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technix:

--- Quote from: T3sl4co1l on May 03, 2016, 12:39:05 am ---snip

--- End quote ---

Does this seem right? Can it handle a wildly varying output?
T3sl4co1l:
Well, obviously it's not designed for it (R14 only covers a ~20% range -- which is more than enough trimming for the application, which is good), but you still have the problem that C14's voltage will follow C17's voltage. 

Why forward converter this time?

This brings other changes:
- IC3 --> UC3844 (set RTCT oscillator for twice the desired switching frequency)
- D6 --> ultrafast (or use RCD clamp snubber instead of primary reaction winding)
- R29 too large?
- D7, D8 will see over 40V peaks as shown, use UF4004's or whatever
- C16, R30 won't do anything as shown; they aren't really important or necessary for this type of circuit anyway.  C15-D6 serves the purpose of Vpk clamp snubber.
- C15 can probably be smaller (10n?), but be careful of resonance against transformer leakage
- R18 is too small for TL431 bias current (min 1mA)
- Remove R19, C10
- R17 --> 100 ohms or so
- Note that any additional load on +18V has to start up only after IC3 starts up.  Q2 should probably be repurposed as a switch gated by aux power.
- In general, L2 and L3 won't carry proportional currents or similar voltages.  L2 also sees 2.5 times more voltage.  C14 will probably develop around 40V, but be highly variable as well.

In this case, it would be acceptable to use a diode (with a current limiting resistor) to obtain aux power.  Remove D7 and replace L2 with 1-10 ohms. Reduce aux turns to suit.

This way, the supply (which is +400V stable, though with AC line ripple) appears across the primary while the transistor is on; the same voltage appears on the windings, divided by turns ratio.  A secondary of 4 turns would give about 16V peak, which will give somewhat less VDC after half wave rectification in series with a current limiting resistor.

The main secondary is probably fine, but expect to change L3 and C17 as needed.  And, in turn, R15 and C8.

Tim
technix:

--- Quote from: T3sl4co1l on May 04, 2016, 02:40:25 pm ---Well, obviously it's not designed for it (R14 only covers a ~20% range -- which is more than enough trimming for the application, which is good), but you still have the problem that C14's voltage will follow C17's voltage. 

Why forward converter this time?

This brings other changes:
- IC3 --> UC3844 (set RTCT oscillator for twice the desired switching frequency)
- D6 --> ultrafast (or use RCD clamp snubber instead of primary reaction winding)
- R29 too large?
- D7, D8 will see over 40V peaks as shown, use UF4004's or whatever
- C16, R30 won't do anything as shown; they aren't really important or necessary for this type of circuit anyway.  C15-D6 serves the purpose of Vpk clamp snubber.
- C15 can probably be smaller (10n?), but be careful of resonance against transformer leakage
- R18 is too small for TL431 bias current (min 1mA)
- Remove R19, C10
- R17 --> 100 ohms or so
- Note that any additional load on +18V has to start up only after IC3 starts up.  Q2 should probably be repurposed as a switch gated by aux power.
- In general, L2 and L3 won't carry proportional currents or similar voltages.  L2 also sees 2.5 times more voltage.  C14 will probably develop around 40V, but be highly variable as well.

In this case, it would be acceptable to use a diode (with a current limiting resistor) to obtain aux power.  Remove D7 and replace L2 with 1-10 ohms. Reduce aux turns to suit.

This way, the supply (which is +400V stable, though with AC line ripple) appears across the primary while the transistor is on; the same voltage appears on the windings, divided by turns ratio.  A secondary of 4 turns would give about 16V peak, which will give somewhat less VDC after half wave rectification in series with a current limiting resistor.

The main secondary is probably fine, but expect to change L3 and C17 as needed.  And, in turn, R15 and C8.

Tim

--- End quote ---

Forward topology: load varies a lot (200mA or so when idle with only the backlight of the LCD on, 3A when all systems go and charging an iPad Pro at full current). Flyback won't take that load variation too well. And the transformer design for flyback bites me hard.
R14: I just need to trim the backplane LV rail somewhere around 5V, so this range is okay. 5.3V is usually a good value for USB charger ports and that is why I trim for that. For primary output strips I use external error amps and ADC for voltage (and current) settings.
IC3: I will probably spin for the cheaper of UC3844 and UC3845. Won't hurt, I believe.
D6: US1M is SMT version of UF4007 (I default to SMT whenever possible.) So already ultrafast. RCD clamp have a lower efficiency as energy is burned in the clamp, but the reset winding returns the energy to the mains filter cap.
R29: 1V/(40W/400V*4) = 2.5ohm. Yup, should have spun for a 2.5 ohm one here.
D7, D8: Either SS16 or US1M should work. Same package anyway, so draw SS16 in but leave headroom for the higher voltage drop from US1M. Install whichever is cheaper that week and it will work.
C10, C16, R19, R30: I'll just leave the pads there, unpopulated (mark those as DNP.) Fairchild's AN included those though, so just as a safe move.
C15: I though about that... Going back to 10n
R17, R18: 100 ohms for each of them?
Q2: The PFC chip also shares this +18V rail, so does the output strips. So maybe I can leave D7 unpopulated and short out pads for L2 but keep the assisted Zener in there.

For switching other users of +18V I would use a MOSFET for that. Again, lower loss.

I am way oversizing L3 and C17 (as to keep the output ripple within the acceptable range for MCU and USB I only need 1/4 the capacitance of C17 and 2/3 of the inductance of L3) to keep the ripple down. I have a habit of doing this - in my previous designs even a MC34063-based boost SMPS that drives only one op amp and one MOSFET gate (linearly) have a specified 10mV maximum ripple on the 15V output - just a few hundreds of ppm.
T3sl4co1l:
You don't have the option of reducing ripple by using overly large inductors: you're limited by the controller.  A peak mode controller goes unstable (chaotic, in fact) in CCM.  You don't have any slope compensation shown, but that only extends the operating range some, it doesn't fix it.

This is the main reason why peak current mode controllers are used for smaller applications, like < 100W.

Tim
technix:

--- Quote from: T3sl4co1l on May 06, 2016, 02:43:12 pm ---You don't have the option of reducing ripple by using overly large inductors: you're limited by the controller.  A peak mode controller goes unstable (chaotic, in fact) in CCM.  You don't have any slope compensation shown, but that only extends the operating range some, it doesn't fix it.

This is the main reason why peak current mode controllers are used for smaller applications, like < 100W.

Tim

--- End quote ---

Auxiliary rail maxes out at about 50W. NBD.
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