Picoammeter Design

[Cerebus]

Yes? So? Doesn't allow them to escape physics.

If you try to balance the currents by simply matching the bias voltages you will always arrive at the conclusion that it's best to minimize the bias voltages period ... in which case matching goes out of the window. You could increase the bias voltages to get a better relative match, but the absolute error stays the same, so it gets you nothing.

I just don't see a cheap way to create matched bias currents in the ESD diodes, so I doubt. Do you see one?

PS. AFAIK the bias current for a low leakage diode at 10 mV reverse bias is in the fA range ... so really you need to do nothing more than have a <10 mV offset buffer and create +10/-10 mV guard voltages for the ESD diodes. Once the input escapes the buffer, additional diodes can carry the ESD to the rails. No matching needed. It's the most straightforward architecture.

Sigh. Sorry, I was trying to stop you wandering off into the weeds by (1) saying a working, verifiable commercial design was wrong and can't work like that, (2) going off at a complete tangent because the original you 'kind of quoted' was about shot noise and the basis for calculation thereof, (3) discussing a quote of a data sheet as if it was David's original post. Feel free to grab the wrong end of any stick you like...

[Marco]

saying a working, verifiable commercial design was wrong and can't work like that

Their explanation of how it works is likely simplified to the point of uselessness ... it is likely just bootstrapped.

A more realistic view does allow you to approximate worst case shot noise ... and funnily enough it's just the worst case leakage through a single junction. Because worst case leakage and worst case shot shot noise will have all the bias current going through one ESD diode and not cancelling out.

PS. well I guess that depends how you account for the buffer, regardless their explanation is completely misleading. There is almost certainly not some relatively large current through the ESD diodes cancelling out.

[fcb]

So the ESD structures/diodes on the inputs are held at a guard voltage derived from the input? Presume there are another set of ESD structures to protect the guard drivers?

[Marco]

There's a couple of options, none of which produce large bias currents which cancel out. They produce bias currents caused by offset voltage, which ideally cancel out a bit ... but if not, then not. Cancelling is incidental, keeping them small to begin with the important part.

PS. I think I was overcomplicating things when saying you had to keep the diodes reverse biased, might not matter if they become positive biased a bit.

[NeverDie]

Is this helpful?  TI very recently posted the gerber files for their LMP7721 eval board.  You can find the zip file here: http://e2e.ti.com/support/amplifiers/f/14/t/895571?LMP7721MAEVALMF-NOPB-Gerber-files

[ckocagil]

I built one of these too based on Gyro's design. I tried to stay faithful to the builds here since this is my first experience with a low current instrument (I haven't owned or used one prior to this). Even copied Gyro's box cover

Cap and resistor: 330pF PS cap, 1G 1% resistor (both from Taobao)
Input socket: high voltage BNC socket from Taobao
Cleaning: IPA and brush

Post-cleaning the output started to drift wildly, but it settled after about 10 hours.

Results: nulled to within 10fA, noise is +-10fA within 60 seconds. I'm happy with the result, but I'd still like to know if anyone has a suggestion to achieve a lower noise.

[Gyro]

Looks good, (I'm flattered  ).

I'm not sure how much of the thread you read, but the easiest way to zero adjust the opamp Vos is to connect the input to the -Ve output. Anything remaining when you remove the link is then bias current.

The Johnson (thermal) noise from the 1G resistor is approx. 3uV RMS (fA reading) for the unit's 0.5Hz bandwidth, so around 10fA Pk-Pk. There's no getting around the laws of Physics. It sounds as if you may be a little bit higher than that (20fA P-P?).  I had to screen the end of the BNC connector with a piece of copper tape* to reach the lowest noise (it is very sensitive), lid screwed on too of course.

The only way to reduce noise further would be to reduce the bandwidth (increase the value of the 330pF cap), but of course that would increase the settling time too.


P.S. * Or just hold a coin over the end.

[magic]

The only way to reduce noise further would be to reduce the bandwidth (increase the value of the 330pF cap), but of course that would increase the settling time too.

It has already been discussed here that you can increase SNR by increasing the feedback resistor.

[Gyro]

Johnson noise increases with resistance and I think it is the dominant noise source here. Unless you change the capacitor value too, to restore the same 0.5Hz bandwidth, you are not measuring like for like.

The other, more practical issue though, is that it would increase gain, reducing the maximum current reading possible with the 9V battery, and increase the contribution of the Opamp bias current, which cannot be sensibly nulled out. I took 1G as a sensible compromise (other compromises are available).


Edit: Actually, yes. Thermal noise with resistance isn't a linear relationship, 10G and adjusted for 0.5Hz BW would be quieter. It didn't meet my practical requirements above though.. No, it would still be noisier, just not 10 times as noisy.

[magic]

it would increase gain
it would still be noisier, just not 10 times as noisy.

There you have it, higher SNR. Alex Nikitin or somebody mentioned it long ago.

[SilverSolder]

Switchable gain, e.g. with a relay or even a panel switch?

[Kleinstein]

The current noise from the Johnson noise of the resistor goes down like square root of 1 /R when the resistor is increased. So more resistance helps, but only a little, at the price of a reduced range. With 10 G the Johnson noise would be about 1/3 but the range reduced to 1/10.

There can also be other sources of noise (e.g. the OPs noise - CMOS OPs can have quite some low frequency noise). 10 µV_pp is about the size expected the low frequency range. The effect of voltage noise is reduced linear with a higher resistor.
Another point can be surface charges and dielectric absorption with  isolating material inside the case. This is more like some drifty slowly relaxing background.

Range switching is tricky as the switches can have leakage. So it is possible but not that easy as it looks. Just changng the resistor would also change the time constant, so one would also need a second capacitor.

[mark03]

What if you got rid of the resistor and used a capacitor instead, i.e. an integrator, with some periodic reset mechanism and a sample/hold/display at the output?

[Kleinstein]

Integrate and reset is possible, but it has it's own limitations, e.g. when averaging over more than a discharge period.
It usually also takes some extra HW behind to do the reset and calculate the current from rate of change or at least send the data to a computer. The simple TIA can directly connect to a simple DMM or even analog meter.

edit:
The capacitor charge method gets interesting when the resistor noise gets much larger than the OPs voltage noise. So more as an alternative to resistors larger than some 100 G, not so much for the larger ranges lime 1 nA FS, where the OPs noise is often larger than Johnson noise.

[NeverDie]

In case anyone here is interested, on another thread it seems we've established that Gyro's design can accurately measure picoamps even at single digit picoamp levels:  https://www.eevblog.com/forum/beginners/static-control-requirements-for-picoamp-measurements-using-ucurrent-gold/msg3101836/#msg3101836

Congrats to Gyro for a great design!   

[arivalagan13]

Can anybody tell me how these PAM's can be tested? I mean can we have a simple current source in the picoampere range to test this picoammeter? I want to test a10^9 gain transimpedance amplifier. I want to design a simple current source to test the amplifier. Any suggestions?

Regards
ArM

[Gyro]

Hi,

I've attached the text of the PM that you sent me as it includes more detail than your post, it will hopefully help more people to make suggestions...

I got to know about you from the eevblog picoammeter design thread and thought.
I'm designing a precision Transimpedance Amplifier (TIA) to amplify a current wiggling between few picoamperes to around 100 picoamperes maximum. I built the amplifier with LTC6268 500MHz version.

The problem is when I tried to test the TIA.
I want a picoampere current source to test the TIA in the above said range. Could you please give me a suggestion or an outline on how do I do that in designing a picoampere current source.

Any advice would be much appreciated.

NeverDie's thread in the beginner's section (linked in his last post above) gives a good narrative on how to generate low source currents at DC.

The real reason for quoting your PM text is that you mention that you used the "LTC6268 500MHz version". This makes me think that you are planning some high frequency application. If so, it is going to be much harder to generate suitable low test currents needed for testing due to the influence of stray capacitances.

The LTC6268 seems to have very similar input current DC characteristics to the LMC662 (it bootstraps its input protection diodes too), which seems to be quite remarkable considering its frequency performance. Being able to utilise them at significant frequencies is a potentially a different matter (I notice that their datasheet typical application circuit uses a 20k feedback resistor for a 65MHz TIA, which is a long way from your 10^9 gain).

Maybe you can give us some more detail of your intended application and operating frequency?


P.S. The application information in the LTC6268 datasheet makes interesting reading, particularly with regard to controlling (and making use of) stray capacitances.

[arivalagan13]

Sure. I want to amplify a signal coming out of a Faraday plate detector (part of ion mobility spectrometer). The application does not demand high frequency response. It's speed is mentioned in terms of Full width at half maximum (FWHM) at 0.5 milliseconds (worst case)..this corresponds to an approximate bandwidth of few kilokertz. But, I'm given to design the amplifier for 50kHz at 7 microsecond rise time.

The signal amplitude is between few picoampere to around 100 picoampere. That's about the specs. I hope the information provided is sufficient.

Coming to the TIA design, yes, 10^9 gain is too much of gain. So, I'm going for a two stage design. The first stage amplifier has transimpedance gain of around 100 megohm (will iterate with 250 megohm as well) and the second stage gain is 10 (or 4).

Hope this information helps.

Regards
ArM

[Gyro]

This sounds like an interesting application, I'm afraid only a few fairly random thoughts come to mind at the moment:

- The LTC6268 does not have offset voltage adjustment pins, so there is no way to zero out the Vos contribution. You can still differentiate the relevant  contributions of input bias and Vos however, by the same method of shunting the feedback resistor, measuring the resulting output (which give you the Vos if the opamp at unity gain). Then checking the difference between this and the output with the feedback resistor alone and the input open (but screened) will give you opamp input bias current contribution. Just accept the Vos contribution as a fixed (0.4uV/'C) offset. In India I suspect that the ambient temperature contribution to input bias current may be more of an issue in terms of accuracy.

- Calibration of the TIA at DC is relatively simple, as NeverDie demonstrated. This is really a confirmation though - for practical purposes, the accuracy of the TIA is determined by the accuracy of the value of the feedback resistor.

- AC performance is heavily dependent on the physical implementation of the LTC6268 circuit. As I mentioned, stray capacitances are absolutely critical, and the 'Application Information' section of the Datasheet goes into this in significant detail. LT also have an eval board, the DC2414A, which covers three different PCB layouts.... https://www.digikey.com/catalog/en/partgroup/ltc6268-and-ltc6268-10-demonstration-board/66825 This uses a photodiode as a high frequency current source example. The DC2414A board manual contains even more detailed application information. These documents will probably give you the most specific and relevant information source.

- A 100M feedback resistor still sounds very optimistic given your frequency goals and the significant effects of parasitic capacitance at 7uS rise time.

- Maybe some form of low capacitance C or RC discharge source would give you what you need in terms of AC current calibration source.

[magic]

There is a similar thread in "repair" where somebody is trying to replace a broken 100MΩ amplifier with 30kHz bandwidth. Clearly, it can be done. That being said, I calculated that <0.05pF feedback capacitance was required to get there.

I think calibration at AC would be doable (connect to an AC voltage source through suitable resistor) but one needs to mind parasitic capacitance of the resistor, parasitic capacitance of the TIA and frequency-dependent effective input resistance of the TIA.

[Gyro]

Ah, I missed that one....  https://www.eevblog.com/forum/repair/repairing-a-current-to-voltage-preamplifier/

[RawCode]

Hello

I don't know if i'm going out of topic.

I'm trying to modify this desing making it multi-range, using a push-pull buffer inside the feedback to achieve higher current capability when reading milliAmps.
When i try to use the 1GOhm range, so using the circuit almost as it was design(except for the current buffer) i get a nasty saturated 50Hz output.
do you have any ideas why it happens? I shoud say that i air-wired only the part in the red area in the picture. Can this be the issue?

[Kleinstein]

Good shielding is definitively needed to keep the mains hum out. The switches will add more leakage current. So one would minimize the switching to a minimum, especially with the 1 G resistor active. Not all switches are suitable.

The buffer is not working well unless rare zero threshold fets are used. This can be a problem for the stability as it slows down the buffer at some current.

Even with a buffer the higher current range is not practical, as there would be too much heat at the resistor with more than some 10 mA. Even 1 mA may start to become tricky and requite a reduced range. The other point is that the TIA input may need some "isolation", not to have to much capacitance at the input. This would requite some series resistance, so the useful range in usually limited up to maybe some 100 µA - this would be a current range that does not need the extra buffer, at least not a very strong one.  The 1 mA range would be more like a completely separated TIA.

[Cerebus]

Even with a buffer the higher current range is not practical, as there would be too much heat at the resistor with more than some 10 mA. Even 1 mA may start to become tricky and requite a reduced range. The other point is that the TIA input may need some "isolation", not to have to much capacitance at the input. This would requite some series resistance, so the useful range in usually limited up to maybe some 100 µA - this would be a current range that does not need the extra buffer, at least not a very strong one.  The 1 mA range would be more like a completely separated TIA.

My "rule of thumb" for precision circuits is to keep resistor dissipation down to 10 mW or below. Obviously, like all rules of thumb, this doesn't obviate the need for proper analysis in cases that deserve it, where one will take into account tempco, thermal resistance, etc. etc. As we don't know the later for RawCode's intended components we can't do a proper analysis, but perhaps using rule of thumb would yield some insight.

Assumption, the high current buffer is only really intended to come into play in conjunction with the 1 ohm resistor. Dissipation at 10mA will be 1 * 0.01² mA = 100 uW. So, as long as my rule of thumb holds, we should be golden. Let's see if it does. Assumptions 200ppm resistor tempco (i.e. cheap, non precision resistor), 1/8W resistor (50ºC temp rise for 0.125W, so R_th to air = 400ºC/W). So, 100 uW * 400ºC/W = 0.04ºC self heating => 8 ppm (0.0008%) shift from self heating.

Conclusion, worst case effect of self-heating from a 10mA current on the 1 range is 8 ppm - negligible for a circuit of the likely precision class we'd assign for this class of picoammeter. A 100mA current would generate 10 mW of dissipation, 4ºC self heating and 800ppm (0.08%) shift (Same assumptions as above,  1/8W and 200 ppm/ºC. Obviously could be made better by picking a suitable precision resistor).

[Cerebus]

Hello

I don't know if i'm going out of topic.

I'm trying to modify this desing making it multi-range, using a push-pull buffer inside the feedback to achieve higher current capability when reading milliAmps.
When i try to use the 1GOhm range, so using the circuit almost as it was design(except for the current buffer) i get a nasty saturated 50Hz output.
do you have any ideas why it happens? I shoud say that i air-wired only the part in the red area in the picture. Can this be the issue?

You need to extend your air wiring and shielding to all the high impedance areas. That includes all of the switches that are currently outside your 'red' zone.

I would have designed it with the switches on the low impedance side (i.e. between the buffer output and the range resistors) and kept my highest impedance resistor in circuit at all times and just compensated for it always being in parallel by either different component values or by correcting after the ADC.

Also, in passing, if any of your switches are mechanical (including relays) there is a real risk of not pushing enough current through to overcome 'dry' contacts. If you're using mechanical switches or relays make sure that they are rated for 'dry' switching.