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| SMPS for dormant tube audio amp project |
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| MagicSmoker:
--- Quote from: GK on August 22, 2019, 01:12:14 am ---Will you guys please read my second post in this thread I forced ccm with the low side igbt entirely for the control properties ; constant pwm d/c at all load currents --- End quote --- Hmm, rereading my previous reply I see I didn't word it as clearly as I should have, and I did gloss over your 3rd post where you emphasized you didn't care as much about reducing freewheeling diode losses, even though that is the only real benefit to a synchronous buck. Yes, duty cycle does change from being relatively independent of load current in CCM to being directly proportional to such in DCM, but this is never an issue in the real world because the buck is the easiest switchmode converter to stabilize either way. In contrast, during forced CCM operation of the buck choke (that is, maintaining current flow past the point the inductor current would otherwise go to zero), the synchronous low-side switch allows charge to get sucked out of the output capacitor only to shunt it to ground - basically, then, at near zero load current the upper switch is dumping charge into the capacitor only for the lower switch to suck it back out. So, not just pointless, but counterproductive*. This happens whether the lower switch is a MOSFET or IGBT, but with the latter type the one true benefit of synchronous operation - reduced voltage drop during the freewheeling period - is lost. Furthermore, the benefit of reduced freewheeling losses becomes more marginal as the output voltage approaches the input voltage in the buck; generally speaking, one only bothers with the synchronous buck in low voltage, high step-down ratio applications like, say, supplying ~1V to a CPU from a 12V rail. Neither of these conditions apply to your application. * - EDIT - actually, the converter can actually operate as a boost in deep fCCM mode, depending on how long the synchronous switch is on relative to the period, choke inductance, etc., but let's not complicate things further. |
| GK:
--- Quote from: MagicSmoker on August 22, 2019, 11:46:42 am --- --- Quote from: GK on August 22, 2019, 01:12:14 am ---Will you guys please read my second post in this thread I forced ccm with the low side igbt entirely for the control properties ; constant pwm d/c at all load currents --- End quote --- Hmm, rereading my previous reply I see I didn't word it as clearly as I should have, and I did gloss over your 3rd post where you emphasized you didn't care as much about reducing freewheeling diode losses, even though that is the only real benefit to a synchronous buck. Yes, duty cycle does change from being relatively independent of load current in CCM to being directly proportional to such in DCM, but this is never an issue in the real world because the buck is the easiest switchmode converter to stabilize either way. --- End quote --- The regulator needed to be stable without load. CCM fixes that problem. Having a duty cycle independent of load current also benefits the regulators transient regulation performance in response to sudden and large changes in load current - the control loop doesn't have to slew over a large change in duty cycle. I posed a whole lot of scope shots showing the regulators happiness going from zero load to 320W and then back again. It simply works. And the soft-start control as implemented with the TL494 overrides the control loop until the regulated output voltage is reached - that doesn't work properly in DCM. I can't remember every other design hurdle overcome to arrive at the final design completed almost a decade ago. I'm sure though that there are other ways of skinning a cat. |
| MagicSmoker:
--- Quote from: GK on August 22, 2019, 01:17:51 pm ---The regulator needed to be stable without load. CCM fixes that problem. Having a duty cycle independent of load current also benefits the regulators transient regulation performance in response to sudden and large changes in load current - the control loop doesn't have to slew over a large change in duty cycle. I posed a whole lot of scope shots showing the regulators happiness going from zero load to 320W and then back again. It simply works. And the soft-start control as implemented with the TL494 overrides the control loop until the regulated output voltage is reached - that doesn't work properly in DCM. I can't remember every other design hurdle overcome to arrive at the final design completed almost a decade ago. I'm sure though that there are other ways of skinning a cat. --- End quote --- Well, I can't argue with success, though there are, indeed, many ways to skin the proverbial cat (NB - I find it immensely funny the US and Oz share the same cliche). If you changed the compensation network to Type II (and did more than hand-wave the component values) you'd have no problems with transient response or other misbehavior around the transition from CCM to DCM, and the I haven't used the TL494 in decades - literally, almost 30 years - but there is a comprehensive app note on TI's website that covers how to use it: http://www.ti.com/lit/an/slva001e/slva001e.pdf. I don't recall there being any problems with soft-start and DCM... then again, I don't recall much about the TL494 once I abandoned it and the SG3524 decades ago. |
| T3sl4co1l:
Basically comes down to this: forced CCM is 100% the same control response over all operating conditions, period; whereas, when DCM occurs, the modulator gain at negative command (i.e., negative output current or falling Vout) drops to zero. And so the dominant pole due to the filter cap shifts towards zero. When it falls near the controller's dominant pole, they split and diverge (increased overshoot), and further still, they're pushed into the right half-plane (oscillation). The choice then is: do you tune the controller for very low bandwidth, to keep stability, arbitrarily deep into DCM? (You can't go on forever, because when load current goes to zero, the capacitor pole equals zero: an ideal integrator.) Or do you tune it for CCM operation (prioritizing transient response), and allow the pole to shift into the right half-plane (i.e., it begins oscillating)? Typical operation in this regime is, a short pulse (or burst of pulses), repeating at a frequency much lower than Fsw, and may be repeating erratically (i.e., the error amp's noise is effectively magnified, and shifted up, into switching noise). The repeat rate being lower than Fsw is another way of saying the control loop has a RHP pole, that is about that far from zero. Another aggravating factor is the minimum pulse width, which is usually limited by hysteresis in the PWM comparator, or driver, or of the current sense response, whether propagation delay of a peak current mode control, or bandwidth of an average current mode control. In the latter case, of course, you're more likely to get a burst of pulses. In a lot of applications, the increased ripple doesn't matter -- it's still well within nominal range, because the error amp is otherwise doing its job just fine, and the switching noise is adequately filtered either way. Others, the consistent ripple from CCM may be easier to deal with (or filter) than the erratic DCM response, or the whine of the inductors or capacitors may be too objectionable. CCM supplies still exhibit oscillation at very low output voltages -- minimum pulse width hits you regardless -- but the region out of the total SOA is very much smaller. :) The downside is the constant "stirring" of reactive power, between input filter cap, and output filter choke. The reactive power, divided by the effective Q factor of this loop, equals switching loss. Which by the way, is a handy way to think of switching supplies, that I don't think is mentioned very often; the switching ripple is reactive power, and the total Q factor of everything it flows through, gives losses. I used such an analysis here in regards to core Q: https://www.seventransistorlabs.com/Articles/Core_Loss.html and plotted for a few materials here: https://www.seventransistorlabs.com/Images/Powder_Core_Q.png The takeaway for lossy materials is, you avoid losses by using a small ripple fraction, which is to say using relatively few VARs for a given DC output. That's how you can manage to use a lossy powder core that dissipates 1W, for say 10 VAR switching (Q = 10), and 100W of real output. Alternately if we have very low-loss reactive parts, we can stir some of the device reactives (most often Coss/Cjo) into the switching reactance, and voila, we have a resonant power supply that can run at much higher Fsw, and even achieve higher efficiency, than a conventional (square wave) design would (for the same Fsw). Tim |
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