Electronics > Projects, Designs, and Technical Stuff
SMPS for vacuum tube power amplifiers.(status: back at it)
T3sl4co1l:
DCM vs. CCM, or more broadly the ripple fraction, is determined by the ratio of inductor voltage to inductance and frequency. So, to move towards DCM, use smaller inductor, lower frequency or higher voltage.
Tim
Circlotron:
--- Quote from: T3sl4co1l on April 29, 2019, 12:51:36 am ---A note on peak current mode control: you need to operate in DCM (not CCM), otherwise chaotic behavior results*.
--- End quote ---
CCM with a flyback converter is somewhat tricky. Say you decide to reduce the output so you reduce the mosfet on-time pulse width and this has the effect of *increasing* the converter output pulse width for a number of cycles until the transformer flux ratchets down to a level that matches the desired output. Much harder to stabilise than DCM.
SK_Caterpilar_SK:
--- Quote from: T3sl4co1l on April 30, 2019, 12:26:58 am ---DCM vs. CCM, or more broadly the ripple fraction, is determined by the ratio of inductor voltage to inductance and frequency. So, to move towards DCM, use smaller inductor, lower frequency or higher voltage.
Tim
--- End quote ---
The LM5022 seems to be capable to do both CCM and DCM. It seems to work as DCM up until a certain load. Using the Coilcraft GA3459, which has low inductance or even the GA3460 which has even lower inductance, it might just stay in DCM mode. (GA3459 5uH, GA3460 2,5uH inductance). Also the primary resistance is very low, so this all may just be enought for the controller to stay in DCM.
OR I could use a controller you would recommend. Because I simply cant know every single bit of info existing out there. So yeah I would look into other directions no problem just show me which way :D . Mos common controllers I know is the 384X the TL494,TL497 and these that I just found (LM5022 and others I mentioned in my post somewehre above).
Im also looking at coilcrafts custom transformer request document, maybe try it out, see what the outcome will be. https://www.coilcraft.com/custom_trans.cfm
SK_Caterpilar_SK:
So Im back at this again and now that I see how the powersupply preforms, I want to take it up another level.
Here is what I want to achieve:
Input:
90VAC-265VAC
Output:
6,3VDC 40W (3% load stability, necessary for tube heater life.)
-60VDC biasing output (not much power required so ill just say max 3W, its critical to be stable as much as possible!)
if possible- adjustable output 350V-500VDC 80-100W ( Regulated stable.)
So far I dont know how am I going to achieve this. Im even thinking about using two powersupplies. One for the HV which now could be adjustable, and one for the heaaters 6,3V and the biasing -60V.
Furthermore I dont know what topology to use. Flyback as far I know is only good up until 100W and to add on top of that, I know nothing about switching transformer design so I will have to check out a lot of literature. If someone could point me to some really usefull literature about designing a switchmode transformer and what topology to utilize I would be thankfull.
I have high hopes for you Tim :D hope you have something to send me a link to.
Btw I have been looking at your radio project and damn I like it, I want to build it but with 9pin tubes, I think it will be better in the long run. The only drawback is that its not FM so in my location if you listen to the same radio station over at AM and FM only the news are the same and everything other over the AM band is for old people and the FM band is the mainstream. For that purpose I want to build a pulse counting FM tube radio. I think you already saw it somewhere. Found the thing over here http://diytuberadio.blogspot.com/2011/02/love-grandfather-tube-radios-diy.html
Thanks
Adam.
T3sl4co1l:
Flyback for both is fine. Or forward for heaters, but not for HV because flyback is better there.
I have an FM radio project too that I haven't documented yet: it uses a 6688 RF preamp (double tuned grid and plate circuits, BW = 20MHz), 6J6 single balanced mixer, 6C4 LO. Some of the goals were developing a feel for grid impedances, which are significantly lower than the cold capacitance, when heated and biased. In particular, the 6688 is around 500Ω || 18pF at 100MHz, I think it was. 6J6 is higher impedance (grid to grid), and its cathode input impedance (which is where the LO is introduced) is basically cathode resistance (i.e., ~1/Gm, and that's total Gm of both halves, since they act in parallel for a cathode-LO-input SBM). That constitutes the front end, with an IF output at 33MHz into 50Ω and a final BW of a few MHz.
The IF strip is three 5702s, double tuned for about 200kHz bandwidth and a shitton of gain; except for the first stage which is single tuned (matching 50Ω to the grid), but combined with the front end's IF output network, it's actually triple tuned.
The grid impedances are around 10kΩ plus some pF at 33MHz. Combined with about 5mS transconductance, that's about 50 gain each, but it's more like 100 because the plate load impedance can be matched higher (closer to 20kΩ I think).
Keeping them from oscillating was a bit tricky. The Cg1a for 5702 is relatively high, for high gain at 33MHz. I'm pushing the maximum stable gain, a figure which has to do with the gain and phase shift of the input and output networks, and the feedback between them. There is some neutralization, where the grid network is returned to ground through 100pF, and a small fraction of the plate signal is cap-coupled to that node. The plates also needed shielding, for which I just rolled up some tinned steel, since good luck finding originals? (The 5702s are again mounted in fuse clips, which hold the shield which hold the tube.)
The last 5702 is biased harder, and loaded with about 5kΩ (single tuned) for maximum power output. This is tapped down for a ~1kΩ output impedance to drive the detector. (The reason being, the higher plate impedance maximizes gain and output power, while the lower output impedance gives more drive for the detector.) This drives a 6AL5 detector (half wave doubler configuration, so its average impedance is fairly low), which generates negative detected AM (where applicable). This also serves as ALC, as the grid bypass caps (the 100pF's) are chained together with ~10kΩ ferrite beads and biased with this voltage. For FM, this gives some limiting action already, and for AM, this gives a somewhat logarithmic response, which is fine for detection purposes.
The final IF is further coupled to a 6BN6 sheet-beam discriminator. These are only practical at quite low frequencies (~4MHz IF), due to how the mutual impedances work out, and the width of BCB FM. But IF is 33MHz? For this reason, I have a 2nd LO at 29MHz (a 5744 submini triode), and a diplexing filter network (two different L||C traps, coupling IF to 6BN6 grid, and LO2 to 6BN6 grid, but isolating LO2 from IF final, and vice versa). This gives about 10V of both at the 6BN6 grid, which due to its sharp transfer function, acts as a mixer itself, generating the 4.5MHz IF3 internally as it were.
The 6BN6 is a neat tube, but limited on options, given commercial FM modulation depth, and the beam conductance.
A word about that: the electron beam itself is made of matter, with mass -- electrons. To physically move it around, requires work. That work manifests as a resistance component. The work increases as frequency increases, so the resistance decreases. (In fact, apparently it goes as 1/f^2.) There is also a factor for the beam's space charge working against the driving electrode, which is conservative so manifests as capacitance. It happens that the capacitance is dominant, and for most "high performance" tubes (the best ones of the 40s and 50s, and most from the 60s), the dynamic grid capacitance (at nominal bias) is about 80% of the cold capacitance. The capacitance varies with bias (because of how much beam is passing the grid), and with emission (because the space charge between grid and cathode acts as a spring).
The resistance is much higher than that reactance, so you can still get reasonable tuned Q, and the tube's GBW is still more or less constant with respect to center frequency.
Another aside -- the band width of a tuned amplifier is limited by the source or load resistance, and the minimum reactance of the circuit. It seems all known devices (BJTs, FETs, tubes) have a transconductance-with-capacitors characteristic, with some loss element for the input and output, and none have a fundamental voltage-source-with-inductors characteristic. So, while the same physics applies for both, it seems we only need be concerned by the case with capacitance as the limiting reactance.
Then: the impedance of a tuned network is Zo = 1 / (2*pi*Fc*C), and if C has some fixed minimum, then at a given center frequency Fc, the characteristic impedance Zo has a fixed maximum. Where we have some source resistance Rs or load impedance RL, we have a Q of R / Zo, and a bandwidth BW = Fc / Q. (When Q < 1, it may be more meaningful to think of it as a combined low-pass and high-pass network, and you can consider replacing the inductors with resistors, or with proceeding stage outputs, i.e., making a DC-coupled amp with no LF cutoff (no highpass). In that case, BW is the baseband bandwidth, i.e., the HF cutoff.)
If we have more grid C, we need lower L for the same Fc, and get higher q and lower BW. If we reduce R, we get higher BW, but proportionally lower gain.
So, at a given Fc, an amplifier exhibits constant GBW. Since grid conductance varies with frequency, we should expect GBW to change with Fc, but in practice it's not a big factor.
So, back to the 6BN6 discriminator.
An already-modulated electron beam carries momentum along its path. This manifests as a transconductance between grids. In a tetrode, there is a G_g1g2 which is nonzero -- but the inverse is not typically true, i.e., G_g2g1 is extremely small. This is called a nonreciprocal effect, and happens because the beam acts like a conveyor belt. (Nonreciprocity is very difficult to realize without spending power -- of course, we're talking about an amplifier tube here, so we're doing quite a lot of work, biasing this tube, and this is physically consistent.)
This is used in the 6BN6 to excite the second grid. An impedance is connected to g2, the mutual conductance develops a voltage in it, and that voltage modulates the plate current. When that impedance is a tuned circuit, the phase shift is frequency dependent, and we get a plate current that looks pulse-width modulated with respect to input frequency. This is filtered down to "DC", and hence FM is detected. Slick as hell, isn't it? :)
The downside is, that mutual conductance can only be controlled by cathode current, and only over a modest range. The Q factor of the network needs to be quite low, so that its bandwidth is comparable to the modulation depth -- 200kHz width at 4.5MHz Fc is pretty low, meanwhile the impedance needs to be very high, because the transconductance is very low (in the uS). I ended up using a lot of turns of fine wire (I forget how many, maybe 50-100 of 37AWG on a 9mm form), with a ferrite slug to tune it, to get that little extra impedance out of the network.
The distortion still isn't terribly great. The limiting is quite good -- no fading is apparent, and the response is just as you expect from a modern receiver, as you tune across stations.
The overall structure -- you might notice that there's a lot more tubes in here than a commercial FM receiver, of course partly just because I don't want to have to optimize it that much; but also because I want some more options with this receiver. In the front end, all the tuned networks are pluggable, so they can be switched out for, say, 120MHz (aviation) or 144MHz (amateur 2m) bands. The IF strip is a separate module connected by BNC jacks; if I want a lower or narrower IF (say for AM voice or FM NB), I can construct one. Or add a 50Ω output to the 33MHz final IF, and connect that to a 2nd IF strip, to the same end.
Pictures!
Power supplies -- back on topic in this thread, I use a pair of single-output flyback supplies. They don't seem to generate much noise in the relevant bands, so reception is fine. (They are cleaned up a bit, but I doubt they would pass FCC Part 15, particularly the HV supply which has some nastiness coming from the transformer -- bad windup.)
https://www.seventransistorlabs.com/Images/100WPowerSupplies.jpg
Middle and right, 150-300V 0.5A (set for 150V), and 6.3V 10A.
https://www.seventransistorlabs.com/Images/FMRadio2.jpg Front end, top view (shields off)
https://www.seventransistorlabs.com/Images/FMRadio3.jpg Bottom
https://www.seventransistorlabs.com/Images/FMRadio4.jpg Pluggable networks (FM BCB)
https://www.seventransistorlabs.com/Images/Radio_IF_Filter.png Analysis of 6J6 plate network (front end IF output; left network in above pic). R9 is Ra-a (plate resistance), inductors are air core and coupled as listed, and capacitances of plate (Cga + strays) and trimmers shown separately. Note output is series tuned into 50 ohms -- flexibility in topology is key for good optimization. :)
Cheers!
Tim
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