"The biggest drawback to voltage mode controls is exploding transistors due to unlimited transient response."
That statment makes sense...actually its what im experiencing with my converter. I threw out the 3842 as a no go because I have used it in the past and wasnt really the best ever I seen. I have blown so far 5 fets on my pre pre pre prototype converter. It was getting as hot as the blazes of hell maybe even hotter (MrCarlson). Then that got fixed by puting it on a beefy heatsink and still died after a couple hours. The only transistor that survived all so far is the G30N60. IGBT. And that does heat up but but not so dramatically and id rather blame my PCB for that heating than anything.
You were probably going the wrong direction, leading to positive feedback of frustration.
Bigger transistors have more capacitance = more switching loss, both in the controller/driver and the transistor. IGBTs are also only effective at higher voltages. If this was another low-voltage application, that would be a problem!
You can of course mis-design a current-mode switcher to explode. It's a much more straightforward design process to design one correctly, in comparison to a voltage-mode control that needs several hacks (soft start, current fault, lead-lag compensation with a zero in the filter) to behave acceptably.
(Of those, you're already familiar with soft start; it is what it says. This ramps up PWM slowly, which works when the output is not shorted and when the input is applied suddenly (the usual case). This fails if the output is overloaded or shorted, and this also fails if the input drops momentarily, or a transient load is applied. For that case, a current fault can detect high peak currents (usually with a current transformer and comparator), and retard or stop switching, and maybe restart from a soft-start condition. PITA if your load was expecting stable voltage under transient load, but whatever. The compensation is jargon for an output filter that has ESR as a critical part of its behavior (typically somewhat lossy electrolytic capacitors are used, for their useful ESR), and the error amp has an R+C across both feedback resistors, to give a slight edge in phase margin near cutoff. Effectively, we try to speed up the error amp's response just a little, but also shelf its response just right, to avoid oscillation. Well, until the electrolytics dry up at least...)
Im also not as familiar with the UC3842 as with the TL494 but I guess a look at the datasheet can change a lot of things withing a couple of minutes.
If you need further convincing (and don't have any on hand), build the block diagram yourself, with a comparator (LM393 will do), logic (discrete transistors or a CD4001 or 4011 will do) and driver (complementary emitter follower 2N4401/3 will do). Uh, and oscillator of course. 555 in a pinch, but a... well, anything that can make spikes will do, so among RC oscillators you probably want to use a diode across the timing resistor to get one short edge; 555, CD4093, CD4049, CD40106, etc. will do. So, overall, 4093 probably the best since you can use two for osc, two for RS f-f. Or a spare comparator, that's fine too.
The biggest problem I see with the UC3842 is current function. This may not be a problem with the mains powered PSU, but it sure is with the 12-24 range. There are simply too high currents for a standard resistor to sense the current.
Well not so much current as voltage. Newer controllers will happily sense more like 100mV there (LM5001 comes to mind, although that's a regulator (integrated switch), and I forget what's similar that's a controller (external switch), but that LM5022 is not a bad example). Which makes 10, 20A, or more, practical.
So, two things:
1. You can bias the current-sense node up with a voltage divider from VREF to ISENSE to shunt resistor. This raises the DC voltage at ISENSE, while lowering the AC gain slightly. Instead of a 0-1000mV range, you might have say 700-1000mV range (basically you lop off the bottom 700mV of useful ISENSE range), while only needing, I don't know, 350mV or so of shunt voltage.
1a. You can combine this with slope compensation, which adds DC anyway, but also some AC (namely, from the RtCt pin), which compensates the current loop for operation in CCM (continuous conduction mode -- inductor current doesn't return to zero every cycle). So it reduces the active range on the ISENSE pin (also reducing the shunt voltage drop), and has the benefit of higher inductance (lower core losses).
A note on peak current mode control: you need to operate in DCM (not CCM), otherwise chaotic behavior results*. Slope compensation allows a less-than-100% ripple fraction (i.e., Ip-p / 2 < Idc), although not too much less (50-80% ripple fraction at rated output is typical).
*The underlying reason for this is quite cool: it turns out, a peak current mode controller is an electronic implementation of the Logistic Map, an iterated chaotic function. This function takes a parameter, which as the parameter increases, at first the behavior is nonlinear but reasonable, but suddenly it splits into a multitude of values that cycle between themselves from iteration to iteration (period doubling, limit cycles). The corresponding circuit parameter is, guess what -- inverse ripple fraction. So, keeping the ripple fraction high, prevents chaos. Chaos is undesirable in circuit, mostly because it causes increased ripple and a hissing sound.
2. We don't need to use a resistor. Anything that senses the same current will do. A current transformer is a typical choice:
https://www.seventransistorlabs.com/Images/Mag_Amp_PSU.png(incidentally, speaking of slope compensation -- with the values and components shown, this circuit needs slope compensation! It's an old circuit.)
Note that transformers won't sense DC. We can sense partial DC, as long as it returns to zero every cycle ("pulsed DC"). Which we can ensure here, by using a diode on both sides of the transformer -- well, not explicitly so on the primary, but we can ensure that current is only drawn in pulses, when the transistor is switching, so it works out the same way. A large resistor (1k here) across the CT winding dampens the "reset" pulse, reducing stress on the output diode (FR102).
That looks like this,
https://www.seventransistorlabs.com/Images/Snubber_103Z.jpg(Actually a different module, with a lower ratio CT; same idea, in any case.)
Both of these also show a key improvement, a dV/dt slope snubber. Your attached also shows this (C8, D4, R11). This allows the transistor to turn off, say in 40ns, while the drain voltage rises, say over 80ns (depends on current, because the peak inductor current is charging the capacitor at dV/dt = Ipk / C), greatly reducing turn-off losses. (Turn-on losses aren't usually a big deal, because there's not a lot of inductor current to pull down on at that time.)
The RC value can also be chosen to dampen the free ringdown (when the inductor's current is discharged, before the next cycle begins), saving some more EMI. (The reduced dV/dt already saves some EMI.)
One thing I cant understand at first glance is a lot of the schematics I see on the interwebz like the one I attached dont have direct feedback. Its totally feedbackless? I guess it knows the voltages by the sensing the transient? That would be my only guess. And the losses within the transformer are low enought to not disturb any other rails?
Depends on what's required. Normally, a TL431 and opto is used for feedback. The 3842 is wired as an inverting amp (no compensation), so the opto commands whatever peak current from the 3842 it desires. The internal comparator handles peak current, so that's perfectly fine. The 431 then regulates voltage by controlling that current.
This works fine for one or two outputs, but the problem with multiple outputs is cross-regulation.
Regulating from the aux winding is the same thing: the aux output will be damn accurate, but the rest will be soft and vary up and down with load.
Cross-regulation is driven by leakage inductance between secondaries. Ideally, you want them all very tightly coupled, so they all receive the same flyback voltage, no matter the current draw on each. Practically, 10 or 20% cross-regulation is reasonable. Better is achievable with a carefully designed transformer. Worse is expected from a naively designed transformer...
If you need better cross-regulation, and the outputs are common-ground (as your heater and B+ outputs might be), you can use joint regulation. One TL431 controls the throttle, and its feedback resistor divider is fed by both rails. The regulation is the weighted average (weighted by the ratio of voltages and resistances) of the two outputs.
For a tube amp, this means B+ doesn't fucking explode while the heaters are cold, or conversely, that the heaters just never even begin to heat up while the B+ is completely unloaded. Instead, B+ overshoots some, but not by an insane amount, while the heaters warm up a bit slower than normal, but eventually everything gets there.
Like, I had to do that on this one,
https://www.seventransistorlabs.com/Images/Discrete_Tube_Supply.pnginstead of a single 680k from +100V to TL431, it's actually more like 1Meg, and there's a, I don't know, 47k or something, from 6.2V to TL431. Same 18k from TL431 to GND.
I built two of these; one is in an all-tube set,
https://www.seventransistorlabs.com/Radio_20m/ and one is in my Theremin (which is solid state, with the +100V for varactor bias and optional tube-based timbre circuitry). The latter may be regulated on 6.2V only, I don't remember (which is the main load, so that would be fine).
Now seeing how it kind of works, I can see what the problem could have been with my original design. Tho I was 14 at the timne of designing that 10-24 input to input voltage-60V 50W converter 
. It has a lot of flaws xD , but interestingly I do remeber that the transistor was not heating as much as the diode did, so the fet did give a damn unlike the diode, that was burning hot without a heatsing and even with the heatsink it gets to a modest temp. Im getting off track 
.
Yeah, like I said -- there are fewer things you need to get right in a peak-current-mode controller, and they are easier to get right, but you still have to get them right in the first place.
You probably just didn't know at the time.
Tim
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