Author Topic: FeelTech FY6600 60MHz 2-Ch VCO Function Arbitrary Waveform Signal Generator  (Read 289169 times)

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Offline rhb

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Absolutely fantastic post!

With regard to the frequency response of the factory units, my FY6600 will produce a clean 10 nS pulse at a 10 MHz repetition rate.  I've not looked at it on my LeCroy DDA-125yet, just an Owon XDS2102A I bought for the sake of the 12 bit 500 MSa/S ADC and 20 Mpt buffer depth.  The Owon showed a bit of ringing after the impulse, but until I do some more testing I don't know which is responsible, the DSO or the AWG. The Owon does not have a 50 ohm input, so the thru terminator may be a factor.

I've got a Tek 11801 supposed to arrive Friday.  I received a pair of 12.5 GHz, 23 pS rise time SD-22 sampling heads today.  So with  a bit of luck and hard work I might have some good data on the F***Tech next week. Then I have to contend with how to test things like the SD-22, or worse yet, an SD-32 with a 7 pS rise time.  I started a thread in Metrology on generating <3 pS rise time steps. I swore I'd stop at 3 GHz, but TEA got me anyway.
 
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Online Johnny B Good

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Absolutely fantastic post!

With regard to the frequency response of the factory units, my FY6600 will produce a clean 10 nS pulse at a 10 MHz repetition rate.  I've not looked at it on my LeCroy DDA-125yet, just an Owon XDS2102A I bought for the sake of the 12 bit 500 MSa/S ADC and 20 Mpt buffer depth.  The Owon showed a bit of ringing after the impulse, but until I do some more testing I don't know which is responsible, the DSO or the AWG. The Owon does not have a 50 ohm input, so the thru terminator may be a factor.

I've got a Tek 11801 supposed to arrive Friday.  I received a pair of 12.5 GHz, 23 pS rise time SD-22 sampling heads today.  So with  a bit of luck and hard work I might have some good data on the F***Tech next week. Then I have to contend with how to test things like the SD-22, or worse yet, an SD-32 with a 7 pS rise time.  I started a thread in Metrology on generating <3 pS rise time steps. I swore I'd stop at 3 GHz, but TEA got me anyway.

@rhb

 It's very kind of you to offer your thanks for that post. The more usual response I've had to similar posts made in usenet groups in the past have tended to be of the "TL:DNR" kind, leaving me feeling as though I'd just been casting pearls before swine.  :(

 What I miss in posting into a web forum is usenet's ease of use. However, what I definitely don't miss in posting to this web forum is all the back biting, personality clashes and outrageous displays of personality disorders and the out and out troll postings so typical of usenet.

 It's such a refreshing change not to have one's efforts at making a positive contribution to a discussion being disparaged for its "full and comprehensive coverage". To some extent, I do see their point when in retrospect, I attempt to read out my paragraph length sentences out loud without turning blue from lack of breath.  >:D I guess the folks here must have much longer attention spans (and not have to move their lips when reading).  ;)

 In any case, I post not to seek glowing testimonials for my prose so much as simply to try and contribute something useful or interesting to the discussion so no further thanks are sought nor desired. However, do feel free to pull me up whenever I make any errors or commit omissions. I'm not seeking to lay down ground rules to insulate me from receiving any well deserved criticisms for such errors and/or omissions whenever I might be so careless.  :)

 Having dealt with the "Thank you" for the "Thank you" note, it's back to the business in hand, namely the pulse performance of the FY6600. I'd assumed you'd been referring to the 'impulse' waveform and so tried it with my AWG for comparison. I must have misinterpreted your meaning of the waveform but this is one that's always been problematical for me (presumably because I failed to take note of the much lower upper frequency limit that applies). However, at exactly 2MHz, the impulse waveform disappears. It was only after trying lower frequencies such as 10 and 100 KHz that I realised that there seems to be a bug in that at exact multiples of 100KHz, the impulse disappears completely!  :-//

 Further investigation reveals that it stops producing predictable pulse repetition rates above the 32KHz setting. After checking the specifications given in the user guide appendices, I see that the upper frequency given for pulse waveforms is quoted as 10MHz. However, the minimum pulse width is specified as 20nS which seems to be at the heart of the 32KHz limit for the impulse waveform since it reduces below this limit at a mere 8KHz, growing fatter as the frequency is reduced until at 10Hz, it becomes a 120 microsecond wide impulse, reducing in proportion to the period of the pulse repetition rate before it hits the 20ns limit at 8KHz dropping to 10ns at 15KHz before finally dropping to a reduced amplitude pulse at the limiting width of 8ns at 32KHz after which, the repetition rate just goes to pot.

 I guess that's simply the consequence of trying to provide an impulse waveform from an arbitrary one loaded into the predefined memory slot labelled "Impulse" which makes it one of the least useful of the preloaded waveform options in the machine if you're regularly working with frequencies beyond the audio range. I ended up using the Sinc waveform at 2MHz to trigger my scope when I was comparing the 2MHz PPS signal from my GPS module.

 Having run all those tests on the impulse waveform, it has now become quite clear that you weren't referring to this one but most likely the square waveform, possibly at the 10% duty cycle, assuming you were referring to the width and not the Tr and Tf times which on a 20% duty cycle on mine, gives a clean 20ns wide pulse with Tr and Tf times of 6.8ns according to my Siglent SDS 1202X-E.

 My ambitions to owning fast rise and fall time square wave/pulse generators are rather more modest than yours. For my immediate needs, sub nanosecond will do me for the time being since I'm only interested right now in verifying Siglent's claim of Tr and Tf figures of 1.8ns for the 1202X-E. The FY6600 can only offer at best a mere 4ns or so with a following wind so nowhere near anything to tax the scope.

 The 1ns rating for the NB3N502 clock multiplier chip and the quad clock buffer chips is just quick enough to verify the validity of Siglent's claim which is why I mentioned it. One nanosecond is just the beginning of my possible adventures into the world of picosecond timings.

 Regarding the 3N502 chips, I might have goofed with that choice of clock multiplier since, deep within the pages of the data sheet, there is a reference to a minimum lower frequency limit of just 14MHz on its output. I'd overlooked this fact when seeing the minimum multiplier option of 2 along with a minimum clock input frequency of 2MHz so I could find myself having to multiply the 4MHz PPS output to 20MHz and follow it with a divide by two flip flop chip if it turns out that 10MHz is actually too far beyond the range of the 3n502's lower frequency output limit to work or work reliably. I've got some more testing to do.

 In the meantime, in case you've not read my latest postings to the u-blox GPS module thread, I've managed to successfully divorce the ornamental Patch antenna from the module and repurpose it as a plug in passive antenna, proving that there wasn't anything wrong with it, other than the very strong possibility that it should never have been glued to a groundplane containing some signal carrying tracks in the first place.

 Indeed, it works noticeably better than that 3/4 wave wire antenna I'd made up to test the module whilst awaiting delivery of an active mag mount patch antenna with a 5 metre SMA male plug terminated cable (bargain of the year at a mere 3 quid from a UK supplier apparently, since I couldn't find anything as good for less than a fiver - even passive patch antennas from China were only a quid cheaper!).

 Anyway, now that I've sorted out the mystery of the useless patch antenna, I can procrastinate no longer over the question of the 3N502 chips and 10MHz output. I'm finally going to assemble one into a test jig on a plug in breadboard and face the consequences.

JBG
 
 

Offline rhb

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JBG

As a practical test instrument one of Leo's 40 ps square wave units is the way to go unless you  insist on building it yourself.  Leo is using a Maxim 3949 LED driver with 22-36 ps Tr and Tf.  There is a moral hazard in buying one though.  After you look at the plot of your unit made with Leo's CAS803 using and SD-30 head you may want one for yourself.

Interestingly, my quest for a really fast edge has brought me full circle to the problem of reverberation due to a source embedded in a layer with a perfect reflector at one boundary.  Which also turns out to be the same problem as constructing an ultra wide band antenna using an unbalanced feedline.

I'm using the F***Tech FY6600 V 3.1 "Adj Pulse".  The settings for the attached figures are:

10 MHz frequency
2 V amplitude
10 ns pulse

I've not observed any repetition rate issue other than the BW limitation of the output.  I was able to raise the repetition rate to 60 MHz.  I think it worth noting that CH1 stopped producing output when set to "Adj Pulse".  I then tried CH2, but it does not have "Adj Pulse".  However, when I went back to CH1 and changed the waveform settings the 10 ns pulse returned.

I tried the "Impulse" and it is very poor relative to the Adj Pulse.


I'm using an Instek MSO2204EA with a 50 ohm thru termination taking the outputs with a 2 ft piece of RG-58.

DS0004 & DS0006 are the FY6600 going as fast as it will go.  DS0005 is a Keysight 33622A going as fast as it will go.

DS0004 is with the amplitude at 2 V and DS0006 is with it set to 0.5 V.

DS0007 is my 100 ps pulser from Leo.  Leo-spike is the CSA803 & SD-30 (9 ps Tr, 40 GHz BW) results.

The Keysight is clearly *very* good, but at a list of 60x the F***Tech.  The apparent ringing on Leo's impulse generator is the result of Instek using a zero phase sinc(t) interpolator instead of the correct minimum phase interpolator.   Setting dot mode turns off the interpolation and with infinite persistence you get the result shown in DS0008.

BY digitizing the plots Leo provided and DS0008 one could derive the transfer function of the Instek AFE.

I think it worth noting that for testing scopes, Leo's square wave unit is a better choice.  I have both.  The impulse unit is much more convenient for TDR work.

Have Fun!
Reg
 
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Online Johnny B Good

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JBG

As a practical test instrument one of Leo's 40 ps square wave units is the way to go unless you  insist on building it yourself.  Leo is using a Maxim 3949 LED driver with 22-36 ps Tr and Tf.  There is a moral hazard in buying one though.  After you look at the plot of your unit made with Leo's CAS803 using and SD-30 head you may want one for yourself.

Interestingly, my quest for a really fast edge has brought me full circle to the problem of reverberation due to a source embedded in a layer with a perfect reflector at one boundary.  Which also turns out to be the same problem as constructing an ultra wide band antenna using an unbalanced feedline.

I'm using the F***Tech FY6600 V 3.1 "Adj Pulse".  The settings for the attached figures are:

10 MHz frequency
2 V amplitude
10 ns pulse

I've not observed any repetition rate issue other than the BW limitation of the output.  I was able to raise the repetition rate to 60 MHz.  I think it worth noting that CH1 stopped producing output when set to "Adj Pulse".  I then tried CH2, but it does not have "Adj Pulse".  However, when I went back to CH1 and changed the waveform settings the 10 ns pulse returned.

I tried the "Impulse" and it is very poor relative to the Adj Pulse.


I'm using an Instek MSO2204EA with a 50 ohm thru termination taking the outputs with a 2 ft piece of RG-58.

DS0004 & DS0006 are the FY6600 going as fast as it will go.  DS0005 is a Keysight 33622A going as fast as it will go.

DS0004 is with the amplitude at 2 V and DS0006 is with it set to 0.5 V.

DS0007 is my 100 ps pulser from Leo.  Leo-spike is the CSA803 & SD-30 (9 ps Tr, 40 GHz BW) results.

The Keysight is clearly *very* good, but at a list of 60x the F***Tech.  The apparent ringing on Leo's impulse generator is the result of Instek using a zero phase sinc(t) interpolator instead of the correct minimum phase interpolator.   Setting dot mode turns off the interpolation and with infinite persistence you get the result shown in DS0008.

BY digitizing the plots Leo provided and DS0008 one could derive the transfer function of the Instek AFE.

I think it worth noting that for testing scopes, Leo's square wave unit is a better choice.  I have both.  The impulse unit is much more convenient for TDR work.

Have Fun!
Reg

@rhb

 Thanks... I think.  :)

 It took me several minutes to tie the descriptions to the scope trace captures before I could appreciate what they were showing me. Not a criticism, just a reflection on the complexity of describing the results even with "pictures that (are supposed to) say a thousand words" and the fact that I wasn't at my sharpest at five in the morning. I'm a Night Owl but even I have my limits which I all too often exceed.  :(

 I happened to be using CH2 of the Feeltech (it's less of a stretch for my half metre RG58 BNC lead to reach the scope's CH1 input) to trigger the scope against the 2MHz square wave on the PPS line being displayed on CH2 so I too fell foul of the 'missing adj pulse' option until I re-read your experience and repeated those same steps to discover, as you did, that extra option that curiously only exists in the Feeltech's CH1 wave menu.

 I didn't go all the way to 60MHz, just the 10MHz point but I could see no difference in the pulse shape between 10 and 2 MHz so figured I wasn't going to discover anything new at the 60Mz setting. I got similar rise and fall times (just over 4.5 and 4.6 ns) which didn't alter with pulse width (I tried 10 and 20ns - the latter looks like a nice approximation to a square edged pulse on the Siglent). I didn't see any change either in the timings with amplitude settings right up to the 20vp-p limit (I guess I have the THS3491 opamps to thank for that) so set it to 5vp-p for the rest of my tests.

 Rather intriguingly, the GPS module's drop/add of a 20.8333ns pulse from the 48MHz TCXO clock to keep it in sync with GPS time displayed a hiccup effect on its once per half to five second correction interval I'd been observing over the past few days that I've been assessing the module's suitability to directly generate a 10MHz reference even with the help of that 3N502 clock multiplier chip (if it'll function at such a low output frequency) to get rid of the horrendous jitter that arises out of trying to directly divide down to 10MHz from a 48MHz clock.

 Going back to the Sinc waveform eliminated this hiccup effect, taking it back to its normal synchronising behaviour. Presumably, there's something going on with the adj pulse that's introducing this hiccup effect but I haven't had a chance to investigate this oddity any further. As for the impulse wave option, that's barely good enough for use in the audio frequency range being somewhat useless at any frequencies beyond the absolute limit of 32KHz. You can get output in the MHz range but you need to be aware of it's random repetition rate and the fact that it disappears entirely at exact multiples of 100KHz.

 The NEO M8N module I'm experimenting with is endowed with a 48MHz TCXO rather than a simple XO but it's not adjusted in frequency to keep it locked to the GPS clock signal as I'd originally and rather naively thought had been the case. It's quite clearly left to run at whatever frequency the TC happens to adjust it to and the control is merely a matter of dropping or adding an extra cycle to keep the PPS synced to GPS time regardless of the chosen frequency it's been programmed to run at.

 The lower the frequency, the smaller these phase shift adjustment steps become. I programmed the PPS to 12MHz (another jitter free division ratio) and saw the expected 90 degree jumps, confirming my estimated 15 degree jumps at 2MHz (I plan to multiply this back up to a jitter free 10MHz with the 3N502 chip - or failing that, 20MHz followed by a divide by two flip flop).

 Depending on the time constant used in the 3N502's PLL circuit, this might give me a sufficiently sanitised 10MHz clock reference suitable for use by T&M and communications kit, possibly helped by filtering to a pure sine wave output. This is something else for me to investigate once I've steeled myself to set up the clock multiplier chip add-on.

 I suspect the only way to overcome these phase shift jumps will be to throw a VCOCXO module into the mix but I'd like to justify this expense for myself before shelling out on one. I don't mind the expense of a component when I know without any shadow of a doubt why the component in question is absolutely indispensable to the whole project.

 The mention of my GPSDO project above might look like 'topic drift' but it is germane to my plans to modify the FY6600 to accept an external 10MHz frequency reference as per Arthur Dent's contribution to this thread over a year ago. At least one of the three 3N502 chips I bought will be getting put to good use in this planned modification.  :)

 Obviously, there's no point in adding a 10MHz external clock option without there being a realistic chance of acquiring or fabricating a GPSDO to drive it. I could run the two jobs side by side but I'd prefer to get the GPSDO sorted out (if not fully completed) first before attacking the FY6600 with yet another modding effort. As dramatic an improvement to the frequency stability and accuracy as I've made to my FY6600 with that 50MHz 0.1ppm oscillator board, that's just been a taster of what can be achieved with a GPSDO reference.

 That basic oscillator upgrade had allowed me to run tests with the current GPS module that would have been virtually impossible with the original oscillator chip which had turned the business of trying to compare frequencies with the 'scope into a game of "Chase Will 'o the Whisp". As good as it now is, I just want the even better frequency accuracy and stability that only a GPSDO can provide.

 I completely understand your own TEA issues. I might think I'm not going to be going to the same "extreme" right now but, after reading, back in November, Arthur Dent's OCXO mod with the intention of adding a 10MHz external reference socket and thinking it was rather an OTT modification and electing to go the cheaper TCXO route instead yet now finding myself going to the same "extreme" of adding an external 10MHz reference socket, I'm not quite so sure I'm as immune to TEA as I think I am after all. :-\

JBG
 

Offline rhb

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There were 3 signal sources,  The 6600, the 33622A and Leo's 100 ps impulse.  The DSO sinc(f) issue is *only* visible on the 100 ps signal when the DSO is sampling at 1/10th of the pulse duration.

I have one of Leo's two  channel GPSDOs which has been extremely satisfactory.  The single channel version is cheaper.

I bought a 6 output 10 MHz GPSDO from Roadrunner, but have not yet gotten around to testing it.
 

Offline TheDefpom

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In case it is helpful here is my review of this unit, I didn't see it posted in here already.

 
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Online Johnny B Good

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Thank you Johnny B good for the picture.

Where did you put the minus of the TCXO power supply TCXO (it's seems to go the the earth) and why there resistor (and value)?

Have you made any changes on the power supply (except the ferrite)?

 Hi Gege34,

 I used the co-ax braid for the ground return (-ve) to avoid ground loops. The white wire is the 5v feed picked up from the +5v pin on one of that group of regulator chips. The resistors you can see are a bunch of three 33K resistors in parallel using the -ve screw terminal on the module as a convenient connection point with the other end of the resistors wired to the safety earth pin on the mains socket (it knocks the 90v leakage right down to just under half a volt and neatly avoids mains earth loop issues.

 As far as the smpsu board goes, I've lost count of the number of times I've had it out on the bench for the various modifications I've tried on it. However, the first mod was to wire a 47K across the lower half of the voltage feedback potentiometer formed by the two 10K resistors between the +5v and the 0v rail to boost the 5v to 5.5v in order to raise the 11.5 volts on the "12 volt" rails to somewhere around the 12.7v mark. This wasn't quite enough, even after replacing the weedy rectifier diodes with 20A dual shotky rectifier diodes so I ended up modifying the transformer to add an two extra turns on each end of the 24v centre tapped winding feeding the 12v rails. This raised the voltage to somewhere in the region of 13.7v or so.

 In hindsight, I should have just upgraded all three rectifiers and then added a single turn overwinding to the transformer to buck the 5v winding, forcing the switching IC to compensate and so neatly increase the 12v rails without altering the 5v rail voltage and avoid modifying the transformer since there is ample room to thread (three or four lots in parallel of) the single turn of wire around the existing windings without having to remove the transformer from the PCB as I'd had to in order to to access the ends of the 12v windings in order to unsolder them from their solder tags so as to extend each winding another two turns each.

 Connecting this one turn buck winding is just a matter of lifting the anode end of the 5v diode from the board and connecting the ends in series with the diode's anode and the vacated hole on the PCB. If you connect it series aiding on your first attempt (a matter of dumb luck), all that will happen is that the 12v lines will drop in voltage, leaving the 5v untouched which simply means you need to flip the one turn winding connections round to buck the 5v, forcing the 12v rails to increase in voltage.

[EDIT 2019-03-31]

 I finally got round to undoing the original transformer mod yesterday afternoon to try out the single turn winding to buck the 5v secondary's output voltage as per the above. Not surprisingly, I had to reverse the connections to make it buck the 5v secondary.

 It worked just as I'd hoped it would. However, when it was connected 'aiding' the 5v didn't get above 3.96v (I suspect maybe due to too large a smoothing cap on the 5v rail) leaving the 12v rails  registering just over 8 volts each. Undismayed, I reversed the connections and tried again. getting +4.94v (I'd removed the 47K voltage boosting resistor), -12.98v and +12.83v. I wired in a 200K voltage boost resistor which then gave me +5.07, -13.29 and +13.13 volts which seemed to be close enough to the optimum. I didn't want to go much above the 5v mark in order to reduce the dissipation in those three very hot running LDOs on the main board and the 3.3v LDO on the 50MHz 100ppb TCXO oscillator board itself.

[END_EDIT]

====snip====


 This is just a follow up on the latest version of my PSU modification exercises showing some pictures for the benefit of anyone else who may be contemplating a similar modification. I wish I'd had this "Brilliant Idea"(tm) before I blundered in with my not quite so clever idea of adding two extra turns onto the 12v windings - it would have simplified the whole process and saved a lot of solder into the bargain. Never mind, better late than never. I guess the electronic version of the saying, "Measure twice, cut once!" must be "Think Twice, solder once!". :-[

 The first picture is a view of the overly heavy gauge wire used to overwind four single turn windings in parallel, I could have (and should have) used much thinner wire. It would have eased the job significantly but the temptation to fill that vast clearance around the bobbin overcame common good sense. :(   If I'd used thinner wire, the job of reversing the connections would have been a lot easier (I'd had to unsolder the transformer yet again just to reverse and preform the tails to go back into the hole on the board and line up against the lifted out diode lead).

 The ideal way to form this additional one turn winding is to take suitabley gauged insulated wire (in hindsight some 6 to 8 pieces of much thinner wire) long enough to let you strip only the starting ends with ample to spare to allow them to be trimmed to length and stripped once all the ends of each turn have been twisted tightly together to hold the turns in place to create a low leakage inductance link to the diode and the vacated hole in the PCB. The already stripped ends, identifying the start of each turn can then be picked out and further twisted together and tinned ready for the final connection, allowing the untrimmed ends to be likewise gathered together, trimmed , stripped, twisted together and tinned ready to make the other remaining final connection.

 Provided you've chosen a suitably flexible wire gauge, in the event that the connections need to be reversed, this final corrective task should be fairly trivial with no need to remove the transformer at this or any other stage of this modification. As you can see, I didn't choose my wire gauge wisely, making a rod for my own back when Murphy pounced.  :-[

 The second picture is an attempt at showing the connections to the PCB and the lifted up diode lead, showing just how little clearance there was between the solder blobbed PCB wire and the diode lead. I'f I'd spotted this blob of solder at the time, I would have inverted the board and sweated out that surplus solder to increase the gap (just as well it's only a matter of 10 or 15 volt peaks rather than the 350 v peaks of 240v mains).

 The last picture is essentially just a wider view to provide context (it also shows where I added the 200K resistor to lift the +4.94v to 5.07v to give a little extra boost to the 12v rails (now -13.29 and +13.13 volts with a very light loading). Subsequent testing, driving both channels into 50 ohm loads at the 20v p-p setting at 20MHz reveals a +1.8 and -1.9 volt dc offset range before any hint of clipping starts to appear on the peaks of the sine wave, a vast improvement over the original setup which had zero dc offset tolerance under the same limiting conditions.

 Also worth noting is the 1.2W reduction in energy consumption when each channel is driving a 50 ohm load at the 20v p-p setting with sine waves at 20MHz. Previously, I'd witness a power reading of 9.9 to 10 watts under these conditions whereas now it only tops out at 8.7 watts.The only way to top these consumption figures is to select the square waveform option which increases demand by another half watt.

 This final modification is in addition to the diode upgrade to get rid of the totally unsuited originals. The pair of back to back T220 devices visible in the second and in the background of the last picture are the 20A 45v PRV rated dual shotky low forward volt drop high speed rectifier diodes I used to replace the 12v rail diodes. The 5v rail diode in the foreground of the last image is a 3A Shotky rectifier diode upgrade on the original 1A rated "fast" diode it had been cursed with.

 The diode upgrades not only provide a voltage boost on the 12v rails (nearly an extra half volt) they also reduce the losses in the PSU board contributing to the overall reduction of 1.2W (about 12%) which might seem trivial but anything that reduces both thermal and electrical stress in such a compact piece of test gear is both a very useful and much welcomed side effect in this case.

HTH & HAND  :)

JBG
« Last Edit: May 09, 2019, 04:47:54 pm by Johnny B Good »
 
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Offline JBeale

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I notice that www.feeltech.net has for the past few days at least provided the less than useful response:
Code: [Select]
HTTP Error 502
Bad gateway
76.115.100.230/6bedf6a
2019-04-10 00:26:25
 

Offline bugi

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I notice that www.feeltech.net has for the past few days at least provided the less than useful response:
Code: [Select]
HTTP Error 502
Bad gateway
76.115.100.230/6bedf6a
2019-04-10 00:26:25
Seems to work for me right now. At least it shows something I'd expect, instead of 502.
 

Offline DC1MC

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http://en.feeltech.net/ works (still 85% in Chinese  :palm:).

 Chers,
 DC1MC
 

Offline wasyoungonce

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FeelTech FY6600 60MHz 2-Ch VCO Function Arbitrary Waveform Signal Generator
« Reply #1835 on: April 13, 2019, 12:29:59 pm »
Quick stupid question.  Can the fy6600 fy6800 or dy5300 output dual encoder type pulses to simulate a dual channel encoder. 

Let’s say ttl pulses ch a leading b by 90degrees or b leading a.

I wanted to simulate an encoder that seperates the pulses by 90 degrees but I’d also need to alter pulse period and keep same phase split.

I know I could do the pulses but not sure about locking the phase split relationship

????



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Online Miti

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Yes, it can. You set the wave to CMOS, phase to 90 or 270 degree then press SYNC. Now if you press CH1 > FREQ, you can adjust the frequency on both channels simultaneously and the phase will stay the same. Same procedure for DUTY.
That big spark at power up was by design!
 
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Offline wasyoungonce

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Excellent thanks miti.... I’ll read all the thread to get a full picture but as said need to simulate encoder channels ....plus other uses.
I'd forget my Head if it wasn't screwed on!
 

Offline soundtec

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I performed a few audio tests on my 66' and 68' hundred.
I used a program called 'visual analyser' ,which does Scope FFT and spectrum , thd+noise etc. 'Room Eq Wizard' is another software that can do this ,it will even list the % points of the individual Harmonics 2,3,4,5,6th  , for this test I went with 'VA' for simplicity of setup ,and I felt thd+ noise was adequate for the purpose of the test.

What I noticed was , well what we'd always known , that certain spot frequencies along the way , 3.5khz being one, perform very  much better in terms of THD than other frequencies , it looked like a comb filter effect. as I went up or down the frequency range , the distortion rises and falls in a fairly predictable manner , across most of the middle audio frequencies it reads within spec of .5% ,but at fairly regular intervals along the way it drops to a very much lower number like .002% or even less. The effect of amplitude on distortion wasnt nearly as big a concern as I had thought it might be either ,as expected at very low output levels a higher contribution of noise pushes up the thd+n numbers , the usb sound card input tops out somewhere well down below full output of the gen ,harmonic spikes are visible the moment the sound interface is over driven .

Now supposing we want to make a very low distortion audio 'frequency stepped' test tone on the fy 6600 , by choosing the individual frequencies of the stepped waveform from the bands of sweet spots across the frequency range we could do very much better than .5%thd  of the spec . 
I should have plotted the frequencies  , its obviously a function of DSP clocking frequency ,there must be at least 50 points across the band 20hz -20khz , its easily visible on an FFT though , when you hit the sweet spot all forms of noise and distortion dramatically drop , as much as 20-30 db over worst case .

Hopefully someone  will be able to confirm my results , 





 
« Last Edit: April 14, 2019, 03:12:51 pm by soundtec »
 

Offline Cliff Matthews

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..I used a program called 'visual analyser' ,which does Scope FFT and spectrum , thd+noise etc.
Thanks! Interesting software. http://www.sillanumsoft.org/news.htm
 

Online Johnny B Good

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There were 3 signal sources,  The 6600, the 33622A and Leo's 100 ps impulse.  The DSO sinc(f) issue is *only* visible on the 100 ps signal when the DSO is sampling at 1/10th of the pulse duration.

I have one of Leo's two  channel GPSDOs which has been extremely satisfactory.  The single channel version is cheaper.

I bought a 6 output 10 MHz GPSDO from Roadrunner, but have not yet gotten around to testing it.

 I realise it's been over three weeks since you posted this but I was just reviewing the last page of this forum after checking out ebay for cheap GPSDOs and reading the other EEVBlog BG7TBL forum discussion so your remarks about Leo's GPSDO units rather caught my attention this time round.

 After what I'd read about the BG7TBL units and the fact that they seem to be based on an FLL rather than a PLL (accounting for the almost insignificant 0.15Hz frequency error in these units), I decided to take a fresh look at alternative "cost effective" GPSDOs and landed up on Leo's web site, scrutinising his GPSDO offerings which looked rather nice for the money.

 However, the big turn off for me is the use of a VCTCXO rather than a VCOCXO, so as nice as they looked for the money, they don't quite meet my expectations. They might stay locked to GPS time, unlike those BG7TBL units, but, again unlike the BG7TBL units, won't have the short term stability of a half decent VCOCXO based unit.

 I was only browsing on the off chance of finding a cheap bargain basement unit to tide me over the next fortnight before I get my chance to pick up a  VCOCXO or two at a radioham rally (along with a few other bits 'n' bobs needed for my GPSDO project). I'm seriously thinking of having a go at building that diy-gpsdo-with-arduino-and-1ns-resolution-tic designed by the late Lars Walenius who, rather sadly, passed away on the 28th Dec last year.

 <https://www.eevblog.com/forum/projects/lars-diy-gpsdo-with-arduino-and-1ns-resolution-tic/>

 I'd already (rather speculatively) bought myself an Arduino Nano module when I bought the GPS module, both purchases inspired by Scully's youtube tutorial videos <https://www.youtube.com/watch?v=lbns-FvpzK4>.

 The Lars version will put the Arduino Nano to a less demeaning use than the one it was fated for in the Scullcom GPSDO project which was basically just a cheap 'n' cheerful frequency calibration standard offering no better performance than the bare NEO M8N module I'd bought.

 It's taken me several weeks of pissing around with the GPS module, a sig genny and a 'scope to realise that the heart of any decent GPSDO is its VCOCXO module. That comes first and foremost in the BOM. The rest of the BOM is just the after-market paraphernalia required to bolt whatever choice of GPS that happens to conveniently suit the task (virtually any of the more recent units possessed of a 1PPS output signal) and, optionally, provide "Das Blinken Lights und Switchen"(tm).

 Between my own experiences with TCXOs (the one I'm using in the Feeltech and the one in the u-blox M8N module) and from what I've read on the subject, I know that VCTCXOs aren't in general a good substitute for the VCOCXOs normally used in GPSDOs. That being said, I'm wondering whether you've had a chance to compare that Bodnar unit against any other GPSDOs, possibly that Roadrunner unit you hadn't gotten round to using as of  three weeks ago.

 I know that testing the performance of GPSDOs can be rather time consuming but I reckon it's something worth doing with that Bodnar unit even if it's merely to satisfy idle curiosity about the VCTCXO versus VCOCXO debate in their use as a GPS disciplined oscillator reference source for driving (to keep it on topic) a modified FY6600 for example.  :)

 I just did a quick search for reviews of these Bodnar units and discovered this interesting document file:-

<http://leobodnar.com/files/Informal%20Evaluation%20of%20a%20Leo%20Bodnar%20GPS%20Frequency%20Reference.pdf>

<https://tinyurl.com/y4fgs8oz>

 It gives a fair assessment these TCXO based units with no surprises. A couple of months back, I might have bought the single channel version. I guess my naivety which had led led me to believe I could maybe knock up a poor man's GPSDO version sans the expense of a VCOCXO for a fraction of the price must have put me off buying one.

 TBH, I was seriously considering ordering the £99.99 one just now since, in spite of the compromised phase noise performance, it's still a well made handy little test and calibration unit for very little money but sense prevailed after reading through that report. At the end of the day, it isn't quite up to the standard I'm hoping to ultimately achieve with my own DIY GPSDO project.

 Don't get me wrong, it's a vast improvement over what I'd been hoping to initially achieve but now that "I've seen the light"(tm), I don't want to take any prisoners in my quest to assemble a home brewed GPSDO of my very own manufacture. I've just got to sit tight for the next two weeks in the hope that I'll be able to do a face to face deal at the radioham rally on a used but good quality VCOCXO module or three (and likewise for the other bits needed to glue it all together).

JBG
 

Offline rhb

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OK, this tops it all!!!

I just got the May/June 2019 issue of QEX.  On the front page is an FY6600 which was converted into a WSPR transmitter by the author.  The article references this thread, so hopefully he'll see this and share some details of the project for readers who don't have access to QEX.

He did the TXCO, output amp and mains cord modifications.
 
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Offline 2N3055

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OK, this tops it all!!!

I just got the May/June 2019 issue of QEX.  On the front page is an FY6600 which was converted into a WSPR transmitter by the author.  The article references this thread, so hopefully he'll see this and share some details of the project for readers who don't have access to QEX.

He did the TXCO, output amp and mains cord modifications.

That article is free sample on arrl.org

http://www.arrl.org/files/file/QEX_Next_Issue/May-June2019/Steber.pdf
 
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Offline rhb

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Very cool!
 

Online Miti

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After what I'd read about the BG7TBL units and the fact that they seem to be based on an FLL rather than a PLL (accounting for the almost insignificant 0.15Hz frequency error in these units), I decided to take a fresh look at alternative "cost effective" GPSDOs and landed up on Leo's web site, scrutinising his GPSDO offerings which looked rather nice for the money.

If you're looking for cost effective GPSDO take a look at this
https://www.eevblog.com/forum/projects/my-u-blox-lea-6t-based-gpsdo-(very-scruffy-initial-breadboard-stage)/msg929949/#lastPost

I tested it on a breadboard and it works really well.
That big spark at power up was by design!
 
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Offline rhb

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I bought one of Leo's two output units.  It's *really* nice.  A bit of a hassle to program, but support is good and I'm very happy with it.
 

Online Johnny B Good

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After what I'd read about the BG7TBL units and the fact that they seem to be based on an FLL rather than a PLL (accounting for the almost insignificant 0.15Hz frequency error in these units), I decided to take a fresh look at alternative "cost effective" GPSDOs and landed up on Leo's web site, scrutinising his GPSDO offerings which looked rather nice for the money.

If you're looking for cost effective GPSDO take a look at this
https://www.eevblog.com/forum/projects/my-u-blox-lea-6t-based-gpsdo-(very-scruffy-initial-breadboard-stage)/msg929949/#lastPost

I tested it on a breadboard and it works really well.

 Hi Miti,

 I was just reprising the whole of this thread (Again!) and had reached page 60 where you and the others had been discussing the various firmware update possibilities almost a whole year ago (My! How the time flies by!) when I noticed that you and two other guests were watching this thread so jumped to the last page (74!) to catch up with the latest postings.

 I realised, after looking at the short thread you linked to, started by Gyro on his "My u-blox LEA-6T based fast locking GPSDO [experiment]", that I owed you a vote of thanks. The radioham rally is almost upon me (this coming Sunday the 28th at Blackpool) where I'm hoping to get some face time with actual electronics traders who sell this sort of stuff (OCXOs, ICs and pretty well anything that remotely relates to radio ham activities) and I've added a few more items mentioned by Gyro to my "Shopping List" which includes small 12 to 15 W triple rail +5, +/-12 or 15 volt psu boards (impossible to locate with google/ebay searches) amongst many other parts needed to upgrade my FY6600 and complete my GPSDO project.

 Apologies! I've just realised that I've written yet another 'marathon sentence' that can't be read out aloud without risking asphyxiation.  :-[ I guess the art of packing "everything but the kitchen sink" into one "small sentence" doesn't translate quite so well from that of packing two adults and three small children and luggage into a 1965 Volkswagen Beetle for a 200 mile cross country one week's holiday trip that I still have fond memories of thirty years on. :(

 Anyway, having refreshed my memory in regard of the perceived shortcomings of Feeltech's finest which we had all been obsessing over, I feel that the benefit of (my) hindsight might provide a useful perspective on this long drawn out saga of a thread.

 Putting aside the several pages of commentary needed just to to explain in a nut shell the global geo-politics that's lead us to this point in time where we all seem to be headed for world catastrophe, it's quite clear that Chinese traders using Ebay haven't got the first clue about marketing to the western consumerist societies whose governments will, at the drop of a hat, prosecute wars in 'far away lands' in the pursuit of 'natural resources' to keep their source of income (tax paying consumers) sufficiently satisfied to avoid outright civil unrest and so maintain the status quo of the whole fatally flawed system upon which they rely for a trouble free life of luxury.

 Sorry about mentioning "The Bleedin' Obvious" but I thought it best to remind ourselves of the underlying situation before slagging off our Chinese Suppliers of cheap and surprisingly good (in the main) test gear of which the FY6600 (and its successor, the FY6800) is a shining (fvsvo 'shining') example.

 The most serious complaint we can legitimately raise, is the obvious lack of 'Good Faith" in the Feeltech camp when it comes to customer relations (eg. the lack of interest in putting right the issue of faulty firmware that was 'bricking' the FY6600 units cursed with version 3.0 firmware).

 The rest of this product's shortcomings can be more sympathetically viewed as simply the inevitable consequence of cost cutting trade offs which should be taken almost for granted at the pricing levels involved. A crap product is better than no product at all at this sort of pricing level set to attract an otherwise ill served hobbyist market where such shortcomings can be worked around by a less demanding but more resourceful customer demographic. In this regard, Feeltech's targeting of their 'market' could hardly be bettered (putting aside the obvious shortcoming of customer relations in regard of the V3.0 firmware issue).

 The main obsession appears to have been that of the half mains live 'touch voltage' that exists with any such kit powered from class II smpsus relying upon the mandated EMC bodge of the Y class capacitor to hold conducted common mode switching noise pollution at bay.

 It turns out that the optimum solution to this problem (risk of ESD to a DUT) is simply to use a 3 wire mains cord with a 3 pole mains inlet socket to provide access to the PE to allow a low impedance (10KR) static drain connection to attenuate this half live touch voltage to half a volt or less whilst neatly avoiding the issue of earth loop induced interference at millivolt output levels (both millivolt DC and AC voltage offsets from induction and galvanic/thermocouple effects in the PE earth and mains supply wiring).

 The alternatives of using analogue supplies or mains frequency isolating transformers to eliminate such 'touch voltage' effects without the use of a PE connection are ultimately doomed since it's impossible to completely eliminate capacitive coupling which is responsible for the problem in the first place. The risk may have been reduced (a 1000pF's worth from a Y capacitor reduced to a matter of 50 to 100pF's worth with a small 10 to 20 VA mains transformer - possibly much less with a special isolation transformer, the cost of which being a reduction of efficiency from 95 to 98 percent down to around 70%) but, nevertheless, a risk still remains.

 Pursuing extremely low levels of capacitive coupling to mains voltage interference with special isolation transformers simply to avoid the use of a stiffer three core mains cable and a mains socket upgrade carries the penalty of extra heat dissipation within an already ill ventilated plastic case that's raising component temperatures perilously close to their upper limits for their life ratings (capacitors which may only be good for a few thousand hours at these temperatures versus several tens of thousands of hours by running just a mere 15 to 20 degrees cooler).

 This issue of operating temperatures within the box takes on even greater importance when, for the sake of frequency stability, it is common practice to leave it switched on 24/7 so the avoidance of a low efficiency PSU becomes an ever higher priority requirement.

 The existing smpsu board used by Feeltech turns out to be a much better optimised design than all of the similar three rail smpsu alternatives on offer via Ebay suppliers in that, unlike those offerings, the 12 volt rails have symmetric output capability (the Ebay offerings typically specify 1.5A on the positive rail whilst offering a mere 400mA or even less on the negative rail).

 I've searched and searched Ebay till my eyes bled but haven't (so far) managed to find any dual output (let alone three rail) smpsu boards in the 10 to 20 watt range which offer symmetric current ratings on the 12 or 15 volt rails. In view of the utter simplicity in achieving such symmetry, as demonstrated in the Feeltech design, I'm rather at a loss as to why this should be the case.

 The only shortcomings of the existing PSU board are the lack of screening and filtering and the insufficiency of voltage output on the "12 volt" rails[1]. I've so far managed to address the low 12 volt rail voltage issue by upgrading to proper high speed, low forward volt drop Shotky barrier rectifier diodes and adding a single turn winding to the transformer to buck the 5 volt winding to balance up the voltage distribution between the 5 and 12 volt rails (along with higher value smoothing caps - but only to a limit; too much capacitive loading causes the switching chip to go into an overload state).

 The next step is to add 100nF ceramics across the input caps and mount the board into its own ventilated metal case with room to add additional LPF filtering on the output rails (using large value series input inductors with large value output shunt capacitors to suppress the HF switching ripple noise without inducing an overload response from the switching chip).

 The ventilated metal case being a Faraday shield only requires an "earth connection" to the common ground rail and should avoid a low impedance connection to the PE - being a class II rated PSU, such a PE connection is  not a safety requirement so the risk of ground loop induced interference can remain at bay, courtesy of the 10K static drain connection which nicely serves to suppress the half mains live touch voltage.

 Bypassing this 10K drain resistor with a 100nF capacitor merely reintroduces the risk of earth loop induced HF interference and becomes somewhat of a folly in this case. Mention of which, when it comes to additional mains input filtering, we only need a series common mode inductor between the mains socket and the smpsu mains input live and neutral terminals. Live to neutral capacitors are ok but what we definitely don't require are any additional live and neutral to protective earth capacitive connections. Any such capacitors should have their grounding connection point left disconnected since they'll undo all the good work of the 10KR drain resistor in attenuating unwanted ground loop effects[2].

 Ideally, we should likewise screen the main board if we wish to reduce direct radiation of the higher frequency signals used in testing HF radio gear but this becomes a task of far greater difficulty than that of merely containing the switching hash within the confines of a PSU screening box so is best left alone unless you're the type to see such effort as a challenge to be overcome at any cost.  :) The benefit to cost ratio in filtering and screening the PSU will imo, be a good two orders of magnitude greater than trying to do likewise for the main board so is worth the modest effort involved.

 By all means, replace the existing PSU board with a ready made solution... if you can actually track one down, that is! (good luck with that search for unicorn droppings).

 Turning to the issue of the original frequency accuracy and stability (lack thereof) of the smd XO chip, most of the problem seems to stem from the fact that it was located within just 10mm of the three LDO regulators which were, in my case, running at close to 70 deg C raising the XO chip to 50 deg C according to my IR thermometer. Now that I've reduced the 5v rail from the original boost to 5.49v, in my original effort to get the 12v rails above the 12v mark, back down to 5.07v, those LDO chips are probably now running a degree or three cooler.

 It was no surprise to me when it was reported that just blowing over the XO chip caused a noticeable frequency shift. Indeed, it was this unconscionably high temperature which made me change my original plan to transplant the TCXO from the PCB it came supplied on (cheapest Ebay option for a 0.1ppm 50MHz TCXO) as a direct replacement to using the oscillator board as is and mount it at a jaunty angle over the 50mm 12v fan I had fitted into the base of the case and powered off the 5v rail in order to keep it in the 0.1ppm temperature zone and minimise power up warm up time.

 Prior to that upgrade, trying to compare DIP XO frequency traces against the FY6600's frequency had been a game of "Chase Wil 'o the Wisp" due to that execrable smd XO chip's lack of anything that could be described as 'stability' by even the greatest stretch of the imagination.

 Now, I am able to use it to compare the 8MHz output on the GPS module's PPS line, observing what I now understand to be the infamous "Saw Tooth" corrections being applied to the module's 48MHz TCXO generated clock pulses (20.8333ns jittering at anywhere from 10 per second to once every half minute or so).

 Without such an upgrade to the FY6600, I'd have not been able to observe the actual mechanism by which a typical GPS module disciplines its internal clock to keep its 1PPS output synchronised to GPS time. IOW, this was a modification that was more than well worth my investment in both time and the 16 quid I'd spent on the "clock power board".

 Another modification that's well worth the time and effort (unless you had no intention of generating MHz signals above the 5Vpp mark into high impedance loads[1]) is the replacement of the rather weedy THS3002i dual CFB opamp with a pair of THS3091/3095/3491 CFB opamp chips which will nicely eliminate the gross distortion when trying to produce sine wave outputs into 50 ohm loads at 20Vpp settings (10Vpp into 50 ohms) up to the 20MHz frequency limit before this limit is reduced to 5Vpp at frequencies beyond 20MHz (effectively disabling the use of the THS3002i or its replacements by bypassing it/them for signal levels at or below this 5Vpp limit).

 The only remaining modification that comes to mind that isn't simply a matter of tweaking some preset trimmers on the main board, is that of reworking the 85 ohm attenuator pad into a 50 ohm pad which gets switched into the output by a relay whenever sub 500mV levels are selected. Feeltech are completely aware of this "Skoolboy Howler" since they've compensated for its effect in the High impedance case in the firmware. It cannot, of course, compensate both the high and 50 ohm terminated impedance states since it can only be one or the other. In this case, it's the Hi-Z case since this is unlikely to be noticed by anyone using the generator for audio frequency work since they rarely work with 50 ohm transmission lines and largely deal with feeding signals into 10K or higher impedances.

==========================================================================================

[EDIT 2020-04-16]


 With the benefit of 12 month's worth of hindsight, Feeltech's bean counter knew exactly what that 85 ohm attenuator pad nonsense was all about (reducing manufacturing costs at any price).

 It's obvious now that the design had been based on the industry standard practice of adding a 20dB attenuator and compensating by bumping the signal voltage up by one order of magnitude as you carry on selecting lower output levels in order to hold the effect of quantisation noise at bay for sub 50mV signal levels.

 This was no 'Skoolboy Howler' with some clever firmware patch to compensate for a BoM error to get it to work in the Hi-Z loading case. It was simply an elimination of the more expensive E192 series resistors required to create a 20dB 50 ohm attenuator pad by replacing the 61.2 ohm shunt elements with 100 ohm cheapies and likewise, substituting the 249 ohm series resistor with another 510 ohm cheapie. The values had been selected solely to achieve precisely the same attenuation as the proper 20dB 50 ohm pad it had been designed to work with but only in the Hi-Z case.

 The solution eventually proved to be as simple as "Just fit a 20dB 50 ohm pad, dummy!" after it was discovered that the FY6900 had exactly the same 85 ohm pad and testing the output of the primary opamp driving this attenuator proved that  an order of magnitude step change was being applied at the 500/501mV threshold the attenuator was being switched into or out of the circuit with subsequent testing of my own FY6600 revealing exactly the same order of magnitude step change in voltage associated with the use of a 20dB attenuator just like every single signal/function generator on the market regardless of make.

 The following text (which I've struck through) can therefore safely be ignored as pure speculation.

The lowest impedances they might deal with are likely to be 300 or 600 ohms where the expected discrepancy between 50 ohm source and 300 or 600 ohm sinks will be very close to what they'll observe in this case with an 85 ohm source. The truth of the matter will only come to light if the audio 'engineer' bothers to make the necessary and basic computations to verify the readings.

 This annoying "Skoolboy Howler" with Feeltech's infamous 85 ohm attenuator (the FY6800 remains so afflicted despite the excellent opportunity that Feeltech wasted by not revising the BOM on this newer model's main board), is more likely to become swiftly apparent to those working at RF where the use of terminated transmission lines is more or less obligatory simply to avoid anomalous voltages where cable lengths are often a sizeable fraction of the wavelength of the frequencies involved.

 Indeed, when working with impedance matched circuits, a standard technique for checking whether or not we have a proper impedance match is to observe the expected 6dB drop when connecting to a matched impedance (in this case, 50 ohms). The last thing any RF engineer expects to be doing is to account for an impedance change from 50 to 85 ohms just because the "Amplitude" setting had been reduced to 500mV or lower. Diagnosing problems is difficult enough when the test gear is within specification. Having it change its characteristic impedance just because of a change in output level is an unwanted and unnecessary evil that should never have been there in the first place.

 I've had a go at fixing this annoyance but it turns out to be a lot more difficult to conjure up a 50 ohm impedance attenuator pad that matches the High impedance condition volt drop expected by the firmware bodge. I thought I'd gotten it properly sussed out but after tweaking the temporary trimpots I'd wired up to fine tune my calculated pass element value a lot more than I was expecting to, I'd landed up with a 45 ohm attenuator pad. Close but, as the saying goes, "No Cigar". It was, at the very least, an improvement but I'd like a much better improvement than that but, TBH, by the time I'd finished with all the calculations both pre and post the modification, I'd had more than my fill of attenuator calculations to last me for quite a while.

 I'm still recovering from all that brain bursting effort even now, several weeks on. I figure that if I take a long enough breather, I might succeed with my next attempt at figuring out a new BOM solution. Basically, I'm just working to my main strength which is the art of procrastination. If anyone else fancies having a go at generating a new BOM for these attenuator pads, please, be my guest.

 The key point to this is you have to aim for a 50 ohm attenuator pad which will match what the current pad gives under open circuit loading but only drop 6.02dB when driving a 50 ohm load instead of circa 8.6dB as it currently does. Somehow or other, I'd managed to screw up the calculations and I couldn't come up with consistent results each time I tried. It does look as though a 20dB 50 ohm attenuator pad had been the original design target but the firmware embedded correction to what I presume had been a BOM error means we can't fix it by simply dropping in a 20dB 50 ohm pad.  :(


 This last conclusion where I'd assumed that a firmware kludge had been required to compensate for a "Skoolboy Howler" turned out to be tragically wrong. Dropping in a 20dB 50 ohm pad turned out to be exactly the right solution as I could have determined if only I had thought to test my assumptions by verifying the voltage step change ratio with actual measurements. :palm:

[END_EDIT 2020-04-16]

==========================================================================================

 To summarise, the modifications I think would be considered worth doing by most contributors in this thread (if they haven't done most of them already) are as follows:-

 Upgrade the C8 mains connector to a C6 (allows a thinner cable to be used) or a C14 (tail wags dog effect) in order to connect the zero volt rail to the protective earth connection via a 10KR (or 1KR if 250 to 500mvac is still too much to stomach as an ESD risk) in order to kill off the half mains live voltage (50 to 90vac as measured with a typical DMM) on the BNC shields. Don't directly connect the PE to the zero volt rail - the PSU is still a class II device not requiring any such PE connection. We don't want to make the mistake that Feeltech did with the FY6800's earth connection by introducing an unnecessary grounding loop. This is a modification that everyone will benefit from regardless of their level of interest in frequencies beyond the audio frequency range.

 Install a cooling fan - a small 50x50mm square by 10mm deep 12v fan run off the 5v rail will be more than sufficient to shift the heat out of a box that's not been vented with convective cooling in mind (tilting it on its prop stand actually aggravates the overheating effect!).

 Replace the crappy commodity 50MHz smd XO chip with a half decent TCXO module, preferably, as others have done, by mounting it on a separate board remote from the original's hot spot on the main board. You could choose an OCXO if you're prepared to provide the additional 3 or 4 watts from an extra PSU board (if it's a 5v OCXO, you could utilise the innards of a cheap 2.1A USB wallwart for this task which will run even cooler once extracted from the confines of its unventilated wall plug shaped enclosure).

 If you're planning on an opamp upgrade, then either modify the existing PSU board or else replace it with something better (good luck in finding a suitable replacement though). Apropos of which, the asymmetric nature typical of the current ratings on the +/-12 or 15 volt rails on most of the Ebay offerings could prove a boon if you're planning on fitting a 12v OCXO.  :)

Upgrading the existing dual CFB opamp to a pair of the later spec THS 3091/3095/3491 opamps is a worthwhile modification unless your interests are essentially confined to low frequency audio work or you don't need "Amplitude" settings greater than 5Vpp.

That 85 ohm attenuator annoyance can, of course be worked around with an external 20dB attenuator but if you're regularly working with 50 ohm impedances, then replacing the resistors in that attenuator pad to convert it into a 50 ohm pad will provide a welcome benefit. Unfortunately, I can't offer much help there right now. My previous attempts only got me closer to the nominal 50 ohm than the original 85 ohm impedance match rather than close enough for me to tick it off my to do list.  :(

 Just convert that 85 ohm attenuator into a 20dB 50 ohm pad by using E192 series resistors (or combinations of E24 or E48 series that will give you the required 148 and 61.2 ohm resistance values with some selecting on test if required). The solution is that simple! I'd over-thought the problem but it turned out that the chief beancounter had told their designer to jump and the designer had simply asked "How high?"

 I think that just about covers everything with regard to upgrading this sow's ear into a rayon purse. If I've missed anything, just speak your piece on the matter. I've not mentioned the 4ns jitter on square waves simply because it's inherent to the DDS technology used here and in other far more expensive AWGs, only alleviated in the later yet even more expensive famous brand named test kit. In view of its very low price, I think it's a relatively minor 'defect' we can all (learn to) live with.

JBG

[NOTES]

[1] Unless you're planning on upgrading the THS3002i dual CFB opamp with a pair of THS3091/3095/3491 opamp chips to eliminate distortion of the sine wave output at 20Vpp settings into 50 ohm loads in the frequency range 5 to 20 MHz, there's little point in doing the 12 volt rail voltage boosting mods. Indeed, if you don't select "amplitude" settings above the 5Vpp mark, the CFB opamp chip(s) never even get switched into circuit by the relays. This just leaves you with the business of upgrading the IEC C8 mains inlet socket to a 3 pole socket (C6 or C14 type) to provide a PE termination point for the grounded end of the 10 or 1 K ohm 'drain' resistor to eliminate the ESD risk posed by the Y capacitor.

[2] A perfect common mode choke would be made using bifilar winding but, aside from the mains voltage stresses on the inter-turn winding insulation, the separation of the live and neutral windings which eliminates this risk of insulation breakdown is also utilised to provide leakage inductance which is put to good use to create an effective transverse LPF by the use of additional capacitors across the live and neutral to suppress HF noise ripple voltages riding on top of the mains waveform using the live/neutral pair as a transmission line to reach vulnerable devices and unbalanced parts of the house wiring (eg. lighting switch drops) from where they can re-radiate as interference to wireless devices.

 Most such ready made filters are designed for a LNE setup where a pair of Y caps are wired in series across the L&N X capacitor with the join intended to be connected to the PE, typically via a screening can connection. In this case, such Y capacitor earthing connections are redundant because they now become counterproductive to the need to keep mains half live touch leakage current to a minimum and avoid creating an unwanted earth loop at high frequencies.

 If anyone wants to improve the common mode choke filtering between the PSU board and the mains socket, all that's needed is just a common mode choke or three, similar to the one already on the PSU board (scrapped smpsu wallwarts and the like are a good source). cascaded with X caps across the L & N connections of each section.

 A shielded filtered mains socket where the Y caps are soldered to the PE connected shielding can still be used provided the earth tag is only used to connect to the 10KR drain resistor and not directly to the zero volt rail of the main board.

 Having said all that, the real priority on 'filtering' lies with those LPFs between the DC voltage rails and the main board connections. If you're short on space within the PSU screening box, it's this LP filtering on the DC rails that gets priority. If you've room for both, you can leave the additional common mode choke filtering modification on the mains input side for later once you've verified the effectiveness of the DC filtering mod.
« Last Edit: April 16, 2020, 04:48:34 am by Johnny B Good »
 

Online Johnny B Good

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Voltages and power draw measurement figures for a modified FY6600 on 240v 50Hz mains supply

 As a result of my previous TL:DNR posting, it occurred to me that I didn't have a comprehensive set of test voltages since doing the last PSU modification so I popped the lid off the box (yet again!) and poked at the PSU connector header with my DMM test lead probes whilst twiddling with various settings. I got some interesting results which I think you might be interested in (at least they were interesting enough for me to open the main board circuit diagram pdf to look up the data on those EA2-5 miniature relays and check the THS3491 data sheet).

 For starters, I configured the sig generator for "Maximum Smoke" which simply means driving 50 ohm loads at 20MHz with a "20Vpp" square wave. The 5 volt rail remains at a rock steady 5.06 volts throughout every test and I'd synchronised CH2's wave type, frequency, amplitude, offset, duty and phase settings to CH1's settings so I only need to list the "12 volt rail" voltages which, in this case were -13.03 and +12.87 with a maximum power draw of 9.2W from the mains socket.

For a 20Vpp Sine, the figures were -13.15, +12.97 and 8.7W.

 I didn't waste time on the open circuit, Hi-Z case, preferring to concentrate on finding the least demanding loading state for the PSU which was for a frequency so low it might as well have been zero Hz, aka 10KHz (100 or 1000KHz producing the same results) and a zero voltage output level. Strangely, this isn't the least power consumptive state (although it offered higher "12v rail" voltages of -13.18 and +13.05) with a power consumption level of 5.6W.

 When I increased the amplitude to 0.6v the power consumption dropped to 5.3W as the attenuator relay was released. Curiously, the amplitude setting effects the standby consumption which for me is a minimum of 5W (actually, just a smidgen less on closer inspection) when placed into standby from any amplitude setting greater than 500mV or less than or equal to 5.00V.

 You might well think the relay clicks on switching into standby would be due to any operated relays being released. They may be being released, but only for a brief moment as their previous state is immediately restored as part of the sequence of going into standby mode. What this means is that if you wish to minimise the standby consumption, you need to make sure the amplitude settings are higher than 500mV and less than or equal to 5.00v before pressing the standby power button.

 Of course, the power saving is a rather petite 150mW per relay (300mW maximum) so is unlikely to motivate anyone to worry enough to bother themselves over the amplitude setting whenever they avail themselves of the standby function. However, this behaviour which has an effect on the 12 volt rail voltages, sparked my curiosity as to how much of a load they were presenting to the 5v rail to exhibit such a detectable change in the watt meter readings so I tracked down the data sheet for these relays and found out.

 It turns out that they have a coil resistance of 178 ohms (28mA 144mW loading on the 5 volt supply). Rather spookily, exactly matching the loadings presented by my add on 5 volted 12v cooling fan and that of the 0.1ppm 50MHz TCXO 'oscillator power board' (28mA each) I'd installed to replace the rather shite SMD XO chip that Feeltech's bean counters had cursed the main board with. What a concidence!  ;D

 Although the boost given to the 12v rails by the relay loads on the 5v rail is a bit of a waste when it involves the attenuator relays, it becomes a welcome side effect when the 12dB boost amplifier relays are involved. The 5V rail only has to power a maximum of two relays at any one time since, on a per channel basis, it is an 'either one or the other' or 'neither' usage case, never both at once.

 Discovering the most extreme voltage variation limits of the 12V rails led me to conclude that the maximum values were -13.24 and +13.08 volts for my own particular configuration which event arises out of setting the amplitude to zero volts.

[EDIT] Discovered a slightly higher set of voltages at the 5.1v amplitude setting, -13.27 and +13.10 volts. Presumably due to the absence of any attenuator loading on the THS3491 opamp outputs when the generator's outputs are not connected to anything.

 That represents a peak to peak variation of just 240mV on each rail which is a mere 1.8% on a nominal 13 volt (or a +/-0.9% on a 13.1v mean voltage). That represents a far better voltage regulation than you typically see on the +12v rail with ATX PSUs!

 We might complain about the crapiness of Feeltech's "Special" PSU board but, after undoing the penny pinching effects of their bean counters, it turns out to be an excellent match to its task (pity they didn't get the secondary windings turns ratio better optimised between the 5 and 12 volt windings though - fortunately, quite amenable to a simple single turn buck winding fix on the 5v output).

 Since no one, afaicr, has reported test results after replacing the PSU board with either a shielded high quality low noise smpsu or an analogue PSU to demonstrate the benefit of reduced ripple and noise on the generator's outputs, it's only a guess as to whether installing the original, now improved, PSU board into its own ventilated shielding box with additional LPFs on its outputs will be worth this additional effort. All we can be certain of is that it won't make the situation any worse (provided the job isn't cocked up by well intentioned but misguided improvement attempts such as using a 100nF capacitor to link the zero volt rail to the PE in order to quell the half live mains touch voltage).

 With regard to the minimum voltage requirement on the 12V rails, I took a closer look at the THS3491 data sheet and located the "Headroom" voltage figures. These are listed as 1.2v minimum, 1.5v typical and 1.7v maximum which implies that a +/-12 volt rail will provide a margin somewhere between 800 and 300mV on a 20Vpp output level. In view of the fact that I was able to apply offsets of circa +/- 1.8 volts with a nominal +/-13v supply, it looks like the 1.2v headroom figure  applies in my case.

 I suspect the DC offset adjustment is more likely to be used to zero any offset drift rather than apply a desired DC offset. For anyone needing to actually apply a deliberate DC offset on a 20Vpp signal, it's highly likely they'll be looking for volts rather milivolts of offset. In this case, the simple answer is to invest in a set of cheap alkaline cells and just put it/them in series with the BNC ground return connection (the 10KR drain resistor won't mind unless you're looking to apply significantly more than a 50v dc offset voltage).

 Basically, what I'm saying is that a +/-12 volt supply should suffice to drive a pair of THS3091/3095/3491 opamp chips just as long as it can supply a minimum of 250mA to the opamp rails (worst case scenario 20Vpp square wave outputs plus allowance for the OPA686 opamps which are powered via 5v LDOs fed from the 12v rails).

 Incidentally, that modest voltage imbalance on the 12v rails looks to be a simple mismatch between the transformer windings rather than due to additional loading of the +12v rail as I'd originally supposed. I've scrutinised the main board circuit diagram extremely carefully now and I can't see any additional loading on the +12v rail to account for this minor voltage imbalance.

 I took great care in matching the pair of TO220 dual 20A 45v PRV Schotky rectifier diodes I used, so I'm forced to conclude that there must be some imbalance between the two 12v windings, possibly aggravated by some residual diode forward volt drop mismatch. I rather doubt the two 12v windings were bifilar wound in this case due to the high inter-turn voltage stresses this would impose. They'd have been wound in two separate layers which is likely what accounts for most, if not all, of this modest voltage imbalance even when wound to exactly the same turns count. The mismatch is a fairly consistent 160mV which represents an error of just 1.23%.

 This little PSU board, like everything else in the box, is merely the victim of rampant bean counteritis (and a minor miscalculation of the secondary turns ratio). Feeltech were stuck between a rock and a hard place as far as the provision of DC power was concerned. An overspecified off the shelf unit would have generated excess heat which, in this case of abysmal cooling provision along with overclocked DAC chips, would not have contributed to its reliability in spite of the extra expense involved. It seems (at least after applying some basic anti-bean counteritis remedial repairs), that they found the perfect pebble to wedge between said rock and the hard place - a 'little gem' one might be tempted to say.  :)

 I'm going to measure up the space in my generator for candidate screening boxes to add to my shopping list. If I can't find any suitable ready made three rail smpsus at this Sunday's radioham rally, then by God, I'll get my hands on any suitably sized unit at the cheapest price possible just for an enclosure in which to transplant the Feeltech PSU board and chuck the useless innards onto the scrap pile. There's more than one way to skin a cat!  >:D

 This will probably be one job that'll be left on the back burner to stew for a while but at least any delay won't be for a lack of materials.

JBG
« Last Edit: May 09, 2019, 06:10:19 pm by Johnny B Good »
 

Offline Grillbaer

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Hi there,

Does anybody know how to get a sweep start trigger signal from the FY6800? The "sync out" connector on the backside only outputs a square wave with the current signal frequency, not a sweep start signal.

I did not find anything about it in the manual.

Regards
Holger
 

Offline fisafisa

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Hi
What is the state of the bluepill sofware for Fy6600?

I was thinking to make a version for a STM32 discovery board.
Any chance to get hold of the source code?

Thanks
 


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