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| How much noise floor and other things matter in oscilloscope usability |
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| David Hess:
--- Quote from: Performa01 on December 27, 2021, 11:31:39 am --- --- Quote from: David Hess on December 27, 2021, 03:36:58 am ---The move toward the split-path design was not driven by performance; it was about cost. It happened as soon as low cost monolithic low input current operational amplifiers became available. The cost savings came from replacing the discrete dual matched JFET with a single unselected JFET even though the split-path design requires trimming of the compensation or gain or both. --- End quote --- Well, of course cost might have been a major consideration, even though I cannot see why back then a dual matched FET should have been more expensive than an IC that contains basically the same plus a bunch of additional transistors and other components. Today it’s a different story of course, because these are hard to get and expensive spare parts now, but back in the seventies a dual FET was about as affordable (or rather expensive) as a JFET OpAmp (like LF356) as far as I remember. --- End quote --- Monolithic chips do not require hand grading for precision. The dual matched parts were graded by hand. Note that monolithic dual transistors will not work in this application because of parasitic coupling. Tektronix kept the simpler dual stacked JFET buffer in the trigger circuits where precision was less important. --- Quote ---The discrete differential stages usually did require trimming of the “offset balance”, as far as I remember the old circuit diagrams of up to 300 MHz frontends that did not use a split path topology. Even though your circuit diagram shows three trimmers, I don’t think we’ve seen this in recent designs. Self calibration takes care of the offset error and with modern low tolerance parts in the input and feedback networks the balance between both paths and the transition at the crossover frequency are good enough even without adjustments. --- End quote --- The designs Steve Roach shows (attached below) include automated trimming of the gain of the low frequency path. He briefly mentions noise on page 70 where he discusses the shortcomings of RF MOSFETs. --- Quote --- --- Quote from: David Hess on December 27, 2021, 03:36:58 am --- --- Quote ---Btw, there are folks who have managed to build a balanced version of the split-path input buffer, so you can have this with balanced inputs too. --- End quote --- Haha, I am one of those folks, but it was much lower noise, impedance, and bandwidth for low level DC differential amplification. I extended and improved an existing single ended design to fully differential and it worked perfectly on the first try, which pleasantly surprised me. --- End quote --- Congrats – my hat goes off to you! This was (and still is) true design work, not very common anymore… --- End quote --- The part of it that I really liked was adjusting the frequency breakpoint between the fast and slow path for lowest noise using a sampling DC voltmeter. Low noise was my primary design goal. Then I went back and measured the frequency of the breakpoint and it was exactly where the noise curves of the slow and fast path crossed, right where it should be. --- Quote ---Yes, I’ve immediately noticed that it’s only DC coupled, which means a number of drawbacks, particularly the fact that the input goes open circuit in AC coupled mode, whereas good designs are supposed to have a constant input impedance regardless of the input coupling, or any other settings for that matter. --- End quote --- I do not know that one way is better than the other and oscilloscopes did it that way for decades without problems except where a DC return path was required. AC coupled designs have to sink the gate current somehow which presents its own complications. The reverse engineered Rigol DS1000Z front end that Dave made shows that the input resistance changes when coupling is switched, which has got to be incorrect, but maybe someone could measure it. The big advantage of the AC coupled split-path buffer is that coupling can be switched on the low frequency side with a solid state switch. --- Quote ---The LF-path also doesn’t provide the offset control usually found in DSOs – just because it really is best placed here. But yes, with the low division ratio of the LF input network, the compensation range could not be huge anyway. Nevertheless I have to assume that the offset adjustment is done at a later stage, which means that it actually relies on the usable common mode range of the input buffer – which will of course work to a certain degree because of the relatively high rail voltages of +/- 8.6 V. --- End quote --- The stage following the low impedance attenuator does single ended to differential conversion and that is where offset and position are inserted. Since gain is fixed after that point, the scaling of the position control is fixed, but it was still also intended to operate as a limited range offset control. Adjusting offset at the input buffer in this case would alter the transconductance changing the gain and frequency response, but maybe not enough to matter? Later gain stages include first order correction of bandwidth and gain over temperature. --- Quote ---A maximum sensitivity of 2 mV/div means 16 mVpp full scale. Even 5 mV/div is equivalent to 40 mVpp FS. Since this is hardly enough to drive the plates of a CRT, there has to be a lot of amplification after the programmable attenuator. In a DSO, the ADC would require at the very least several hundred millivolts (but usually up to two volts) full scale for proper operation. This is why integrated PGAs do not only provide attenuation, but amplification as well. Consequently, as the signal needs to be amplified anyway, there’s no need to stop at 2 mV/div. With 20 MHz bandwidth limit the total noise in a proper low noise design can be as low as 20 µVrms, so this should not be a problem for the trace width. --- End quote --- The worst case input signal range at 50 mV/div, where low impedance attenuation is maximum, is +/- 250 millivolts with overrange. The peak-to-peak noise is only apparent in digital storage mode. At the maximum sensitivity of 2 mV/div, the input noise is only just dominates the noise of the following stages. |
| Performa01:
--- Quote from: G0HZU on December 27, 2021, 02:40:05 pm --- --- Quote ---If someone needs a superb instrument for low frequencies, then a Picoscope 4262 is one of the few options – apart from a DSA, that is. The 4262 only has 5 MHz bandwidth, but it is true 16 bits, has an SFDR of >96 dB and a near constant noise density from DC to its upper bandwidth limit. --- End quote --- Yes, I've seen these and there are also some alternatives. Very tempting. At the moment I sometimes use a Tek RSA3408A 8.5GHz RTSA for looking at low frequency stuff. This has a low noise floor and it has the advantage (for me at least) of having a 50 ohm input impedance. The Picoscope should be a bit better although it is limited to a 5MHz BW. The Tek analyser can capture 40MHz but it is only a 14bit system. --- End quote --- I just had a closer look - and sadly my previous statement about near constant noise density isn't true. Even though it clearly is not a split path design and the 1/f corner frequency is significantly lower than for the 500 MHz and 2 GHz scopes that I have here, there is still some significant 1/f noise, slowly starting below some 25 kHz. Well, that's obviously the drawback of an 1 Mohms input impedance, requiring a FET input... Other than the general purpose scopes, there is a major difference between open circuit and 50 ohms termination. Without termination, the noise raises significantly. The noise density stays below 7 nV/sqrt(Hz) at and above 20 kHz, but gets as high as 102 nV/sqrt(Hz) down at 100 Hz. The first two attached screenshots show the noise spectrum at full sample rate up to 100 kHz and at full bandwidth. The noise density is generally higher than in the general purpose scopes (where it is in the range 2-3.5 nV/sqrt(Hz) at and above 1MHz), which might have to do with the higher sensitivity of these scopes. The Picoscope 4262 is limited to 20 mVpp full scale as the most sensitive range. EDIT: Caution! this is for AC coupling with incomplete termination, which results in bad LF performance. Pico_4262_Noise_50_5M_D100k Pico_4262_Noise_50_5M Next comes the noise density graph: EDIT: Caution! this is for AC coupling with incomplete termination, which results in bad LF performance. Pico_4262_ND_50_5M A distortion test at 20 kHz Signal_1V_20kHz And finally a two tone intermodulation test, demonstrating the SFDR (just look at the cursor measurement; the automatic measurement failed because it obviously isn't intelligent enough to operate on the whole trace): Signal_IMD_40mV_20-21kHz EDIT: The noise measurements shown so far did not show the true performance, because they were flawed for two reasons: 1. The input was AC coupled by accident, which of course increases LF-noise significantly. 2. The input had a 50 ohm through terminator fitted, but since this scope is sensitive to the source impedance, an additional 50 ohm end terminator should be used to complete the 50 ohms setup. So I've added the correct measurement results for spectral noise and noise density: Pico4262_Noise_25_5MHz_D50kHz Pico_4262_ND_25_5M |
| mawyatt:
The two tone IMD looks good as one would expect from a "True" 16 bit system. If you don't mind could you do this test at ~1MHz with the Picoscope 4262? BTW one of the reasons almost everything analogish in complex chips is differential is you can't get a good ground reference on-chip for larger size chips. Later when analog type flip ball bond chips became available the on-chip ground reference was better than with traditional wire bonds since these ball bonds could be located within the chip boundaries as required and thus offered a lower ground impedance. Best, |
| G0HZU:
--- Quote ---I just had a closer look - and sadly my previous statement about near constant noise density isn't true. Even though it clearly is not a split path design and the 1/f corner frequency is significantly lower than for the 500 MHz and 2 GHz scopes that I have here, there is still some significant 1/f noise, slowly starting below some 25 kHz. Well, that's obviously the drawback of an 1 Mohms input impedance, requiring a FET input... --- End quote --- Thanks. The Tek3408A RTSA can be very laggy and frustrating to use at times but it is very powerful. The front end is 50 ohms and the noise figure at low frequencies is about 20dB. I've not looked to see how noisy it is below 1kHz but it's bound to get a bit noisier here. |
| Performa01:
--- Quote from: David Hess on December 27, 2021, 03:27:38 pm ---Monolithic chips do not require hand grading for precision. The dual matched parts were graded by hand. Note that monolithic dual transistors will not work in this application because of parasitic coupling. --- End quote --- Thanks for the explanation – this makes sense of course. --- Quote from: David Hess on December 27, 2021, 03:27:38 pm ---The designs Steve Roach shows (attached below) include automated trimming of the gain of the low frequency path. He briefly mentions noise on page 70 where he discusses the shortcomings of RF MOSFETs. --- End quote --- Yes, this well known article is brilliant indeed! Yet we can see its age by looking at the first schematic, figure 7-1: the 50 ohms termination is accomplished by just a resistor, that is connected in parallel to the ordinary high impedance input with its high shunt capacitance – a solution that is barely suitable for scopes with a bandwidth exceeding some 100 MHz. --- Quote from: David Hess on December 27, 2021, 03:27:38 pm ---I do not know that one way is better than the other and oscilloscopes did it that way for decades without problems except where a DC return path was required. AC coupled designs have to sink the gate current somehow which presents its own complications. The reverse engineered Rigol DS1000Z front end that Dave made shows that the input resistance changes when coupling is switched, which has got to be incorrect, but maybe someone could measure it. The big advantage of the AC coupled split-path buffer is that coupling can be switched on the low frequency side with a solid state switch. --- End quote --- It’s been quite some time, but I think I remember that this reverse engineered Rigol schematic has a number of errors in it. Some are more obvious than others. It is a nice means to get an overview, but certainly not suitable to study any circuit details. Well, just because the AC block has been in the input path for a long time, especially when the bandwidth of a scope was rather low, this does not mean that it is a good thing to have to be prepared for unexpected changes in some major characteristics, when operating a switch that basically just alters the frequency response. Consider a high impedance (100 Mohm), x100 high voltage probe connected to 2kV. If you now switch to AC coupling by accident, the input DC-block capacitor will charge up. Current is limited by the probe resistance, but after 10 seconds the capacitor might be charged to about 1.9 kV and this is equivalent to some 60 mJ of energy. So if the (supposedly) 400 volts rated capacitor doesn’t break down (and suffers damage or at least permanent degradation), it will send a potentially destructive pulse of electric energy into the frontend as soon as someone connects a low impedance source to the input after that incident. --- Quote from: David Hess on December 27, 2021, 03:27:38 pm ---Adjusting offset at the input buffer in this case would alter the transconductance changing the gain and frequency response, but maybe not enough to matter? Later gain stages include first order correction of bandwidth and gain over temperature. --- End quote --- Figure 7-3 in your document shows the usual approach where to feed V_offset. In your circuit diagram of the Tek 22xx the offset voltage (delivered from an OpAmp with close to zero output impedance within the LF frequency range) would have to be fed into the lower leg of R98 (after disconnecting it from ground, that is). But with the low division ratio, which clearly is an attempt to keep the LF noise down, there is almost nothing gained, so I can completely understand why it’s done differently in this particular case. EDIT: Sorry, only now i've checked what you mean. Of course, with the transistor output stage the original approach for offset compensation cannot be used. |
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