If you need the channel to be fully differential, you can always just duplicate the circuit - you add some variability to the system, but given the bandwidth you can probably just make sure your gain resistors are tight tolerance and your opamps have low input offset voltage and you'll probably be fine so long as the two are colocated on a board or in the same enclosure. If you need a differential input, you can duplicate the first low gain stage and then send each into a secondary amp to move to single ended and do your final gain stage with that. If you want differential input into the analog discovery for whatever reason but are fine with single ended inputs, you can just have an inverting follower amp after the output stage to get a mirrored signal (though I don't think it's going to promote signal integrity unless you need big time EMF rejection between the output of the preamp and the input of the analog discovery).
Personally, I don't know why you'd need a differential signal for this application, but perhaps your situation can demand it. I would certainly prefer two distinct channels to play with and just go into a differential mode in software, and at these frequencies and signal levels, unless your area is very noisy in terms of EMF (and if it is, you probably need to shield the analog discovery too), single ended should be able to give you plenty of signal integrity and responsiveness within the bandwidth you're looking at and will be cheaper to implement (more choices of parts, cheaper options, fewer parts).
What do you mean by oscilloscope "hook"? (obviously it is not the hook connectors for the probes..)
Getting Rid of Hook: The Hidden PC-Board Capacitance
You probably will not have a problem but it is important where high impedance dividers are used like on the AD2.
8
Is fast overload recovery a requirement? Shunt feedback amplifiers have problems with this.
I read somewhere (cant remember exactly where) that common mode rejection performance is bounded by the tolerance of the matched resistors and even selecting 0.1% resistors does not work wonders in that aspect. But other design aspects would be more important anyway.
Getting Rid of Hook: The Hidden PC-Board Capacitance
You probably will not have a problem but it is important where high impedance dividers are used like on the AD2.
Looks like a complex problem to address. I trust that they dimensioned the variable capacitor sufficiently in the AD2 (5-20pf) to roughly compensate for all these effects. As a side note, AD2s come precalibrated from the factory but it could be that they would also benefit from recalibration every now and then, especially after severe environmental changes or aging.
Fast overload recovery is not a requirement (how fast are we talking here?) But I have always been wondering whether there exists is a distortion free, noise free and "cheap" way to indicate overload with e.g. an led or digital output..
I wonder to what extent hook is still a problem with todays pcb manufacturing methods. Maybe nothing has changed from the 70s but modern literature on this topic seems inexistent.
Also the article mentions teflon standoffs as a way to support critical components whereas they don't mention slot cuts, todays CNCs make them in any shape and width, that is my preferred method of 'isolating' high impedance points.
The article also doesn't mention how the orientation of components in the board my affect performance in some cases, specially in HV circuits, this has to do with the orientation of fibers in the PCB
Beacuse the dielectric strength varies from one axis to another, I’m talking under many tens of kVs, with pcbs operating in transformer oil, the limiting factor becomes not creepage or clearance but the paths inside the pcb through the fiber
I have been investigating options for a high impedance differential preamp for looking at tiny signals in the audio band. My application right now is for probing differential microphone-level signals, somewhere between 5-50 mV RMS. The output would be single ended going into a typical scope 1 meg input.
The Tek AM502 would probably be exactly what I'm looking for but I'd kind of like to make this a DIY learning project and also see if lower noise is achievable.
I've been thinking about using the LSK489 as a differential JFET input pair
The LSK489 would be in front of perhaps the AD8429 instrumentation amp http://www.analog.com/media/en/technical-documentation/data-sheets/AD8429.pdf. 1 Mhz bandwidth would be adequate, but higher would be welcome.
I'm right now trying to figure out the gain and probe matching options to optimize for CMRR and noise. The Tektronix probes with trimmable DC resistance (and capacitance) to match pairs for CMRR like the P6023 or P6135 are nice but they are 10x and it seems like these would really only be useful for larger signals, unless I am mistaken.
Then again the differential amplifier's rti noise decreases with gain, so maybe I am looking at this the wrong way and should go for 10x probes.
There is also the option of simply using a shielded twisted pair and making a diy probe that way.
Beacuse the dielectric strength varies from one axis to another, I’m talking under many tens of kVs, with pcbs operating in transformer oil, the limiting factor becomes not creepage or clearance but the paths inside the pcb through the fiber
To prevent ionization where the dielectric constant falls? I know this is an issue with bubbles in the potting material used in high voltage assemblies.
The LSK489 would be in front of perhaps the AD8429 instrumentation amp
Using the JFET only as an impedance transformer seems like a shame. There are circuits which preserve the noise level of the JFET.
Just throwing this in the pile, maybe it'll be useful to you at some point.
If you want to go all the way down to DC and up to say 10 or 20 Hz, you can use a split path amp. You can have you ac-coupled amp, say 10Hz to 1MHz then, also have something like a LTC1050 handling the DC part, in parallel. Identical gain, of course.
There's something regarding this in App Note 106 fron Linear.
When you post the schematic, also list the test conditions of the LT1102 circuit.
SPICE models do not always model every aspect of a part. The better ones include a list of what is modeled.
Properly DC bias the inputs, not through the diodes.
David I had in mind the -GP version 1N4004GP which is Glass Passivated with 8pF junction capacitance, 5uA reverse current (at 100deg celcius at 400V peak reverse voltage)/(or just 20-50nA at 25deg celcius). Reverse recovery is 2usec though.
Why do you consider the 1N4148 as unsuitable? The worst case leakage current of 50uA is specified at 150 degrees celcius BUT at a more normal 25degrees it is in the region of 20-100nA depending on the (reverse) voltage.
Or do you follow worst case specs because you can expect a sudden junction temperature increase during overvoltage events?
As a side note, no reasonably priced through-hole part can come close to the specs the BAV199 that you mentioned.
And here is another take with an INA111. With similar layout as the previous posts for the LT1102. But it works this time! I will recheck the importing of the LT1102 model..
David I had in mind the -GP version 1N4004GP which is Glass Passivated with 8pF junction capacitance, 5uA reverse current (at 100deg celcius at 400V peak reverse voltage)/(or just 20-50nA at 25deg celcius). Reverse recovery is 2usec though.
But maximum reverse current at room temperature is 5 microamps and reverse current is only weakly related to reverse voltage. The 8 picofarads of capacitance gets doubled with two diodes resulting in a total input capacitance high enough to present problems with probe compensation and bandwidth.
Why do you consider the 1N4148 as unsuitable? The worst case leakage current of 50uA is specified at 150 degrees celcius BUT at a more normal 25degrees it is in the region of 20-100nA depending on the (reverse) voltage.
20 nanoamps into 1 megohm is 20 millivolts and this leakage varies directly with temperature.
Worst case high temperature leakage current into the LT1102 inputs is about 2 nanoamps and leakage current at room temperature is more like 100 picoamps.Or do you follow worst case specs because you can expect a sudden junction temperature increase during overvoltage events?
Leakage from junction temperature increases due to overload is actually a consideration for fast overload recovery.
It is worth mentioning where these relatively high absolute maximum leakage specifications come from. It is expensive to test for low leakage (tester time is charged per second) so a specification like 5uA or 20nA represents what the tester itself was capable of within the time allowed for testing. Where the LT1102 datasheet says +/-60pA maximum at 25C, +/-400pA maximum at 70C, and 15nA maximum at 125C, they are not kidding even though the typical specifications might be 10 times better. 2N3904 base-collector junctions are specified to be 50nA at 25C because of the test itself but are typically more like 10pA and better.As a side note, no reasonably priced through-hole part can come close to the specs the BAV199 that you mentioned.
Dedicated low leakage diodes were never commonly available and technically the BAV199 does not count as low leakage either because it is only tested to be less than 5 nanoamps.
Manufacturers have used small signal transistor base-collector or base-emitter junctions as low leakage diodes for decades. Like the BAV199, they are not tested for low leakage either but demand was never high enough to manufacture a tested low leakage part at an economical price. The user has to qualify or test them themselves which is not difficult.
If you want an inexpensive already tested low leakage diode, then a 2N4117/2N4118/2N4119 low input bias current JFET (10pA at 25C maximum and 1pA typical) is the most economical choice and a lot of manufacturers use low leakage JFETs for exactly this purpose; just tie the drain and source together.
And here is another take with an INA111. With similar layout as the previous posts for the LT1102. But it works this time! I will recheck the importing of the LT1102 model..
That is weird; I was going to suggest that the SPICE was modeling the conductance of the 1N4004GP diode attenuating the input but obviously there is something wrong with the LT1102 model. At zero volts, diode conductance is about 26 mhos/amp. (1)
(1) NIST's Guide for the Use of the International System of Units (SI) refers to the mho as an "unaccepted special name for an SI unit", and indicates that it should be strictly avoided. - Fuck you NIST. I no longer accept any of your advice since you approved deliberately compromised NSA algorithms and standards for cryptography. (2) You are not to be trusted with anything. You are as bad as the FDA. Die in a fire.
(2) Yea, this is just an excuse. I would use mho in place of siemen anyway. But NIST really is no longer to be trusted at least with anything having to do with cryptography and computer security. They can still die in a fire.
If you want an inexpensive already tested low leakage diode, then a 2N4117/2N4118/2N4119 low input bias current JFET (10pA at 25C maximum and 1pA typical) is the most economical choice and a lot of manufacturers use low leakage JFETs for exactly this purpose; just tie the drain and source together.
All this discussion and using a jfet or transistor as diode replacement reminds me of the "biased diode clipping circuit" and the transistor clippers (e.g. BC456 and BC556 NPN/PNP transistors for diode replacement) analyzed in Chapter 24 of Small Signal Audio Design by Douglas Self. There he also mentions that circuits like that have a very high output impedance and thus he always includes an followup op-amp buffer. I wonder whether such a concern is relevant to our case here.
I will certainly try to test the JFET solution though, any pointers to relevant schematics of manufacturers?