1G @ 10hz = 13fA rms resistor noise, approximately? That would be not including current and 1/f noise, correct?
Thanks
1G @ 10hz = 13fA rms resistor noise, approximately? That would be not including current and 1/f noise, correct?
Thanks
If you sample at 10Hz the bandwidth should be about 5Hz and RMS thermal noise current of the resistor would be about 9fA at room temperature. To reduce it you need to use a larger resistance. A 100G resistor would give you ~ 0.9fA RMS noise. Amplifier noise components are added to that figure, though good electrometer-grade opamps add very little.
Cheers
Alex
"wouldn't the thermal noise of the 1G or 10G resistor swamp out the current reading? "
Current noise vsm voltage noise.
The academic advise of using a T networks consisting of loww value resistors here is absolutely the wrong approach. The bigger resistor value, the better.
I made two similar ones before, battery powered, split supply and air wiring at input pin of the opamp. I use 100G and 1T as feedback resistors to achieve lower current noise. Also, I use a piece of twisted PTFE insulated wire pair as the feedback capacitor.
I tried different opamps and different resistors, they are not all good.
Is the top op-amp circuit a constant current generator?
Welcome pigeon,
No it's a rail splitter (basically a voltage follower with the input driven from the supplies by two equal value resistors). It sets the ground reference of the circuit to be 50% of the supply so that the opamp can swing positive and negative. This allows it to respond to both positive and negative input currents.
Welcome pigeon,
No it's a rail splitter (basically a voltage follower with the input driven from the supplies by two equal value resistors). It sets the ground reference of the circuit to be 50% of the supply so that the opamp can swing positive and negative. This allows it to respond to both positive and negative input currents.
Only the resistor and capacitor positions must be swapped - resistor as a feedback and capacitor as a filter to ground.
Welcome pigeon,
No it's a rail splitter (basically a voltage follower with the input driven from the supplies by two equal value resistors). It sets the ground reference of the circuit to be 50% of the supply so that the opamp can swing positive and negative. This allows it to respond to both positive and negative input currents.
Only the resistor and capacitor positions must be swapped - resistor as a feedback and capacitor as a filter to ground.
Eh? It's setting the DC ground so why would you want to only AC couple it to ground? If I remember the circuit correctly, the capacitor you're talking about is there to cut the op amp bandwidth. Or have I misunderstood - you've not exactly made it clear what you're referring to?
That is what I mean.
Look carefully, the 100 ohm and 1M resistors are actually in series going to the inverting input of the amplifier, forming the DC negative feedback, it's a bit confusing because both go to "ground", but ground is the output of that amplifier.
And of course the simpler follower
Ah, now I understand the confusion, pigeon quoted zlymex's post which also contained a (different) schematic.
As Cerebus and iromero point out there needs to be a DC feedback path. The output of the opamp is driving the 'ground' via the 100R to buffer it from capacitance involved. The opamp is sensing the 'ground' voltage via the 1M to establish the DC return path of the voltage follower. The 10n capacitor provides local HF feedback around the opamp for stability.
I found these component values to be needed for stability
in my implementation. The LMC662 was 'twitchy' in driving the ground (low frequency instability). I suspect that it is because 'ground' in my case was the copper groundplane of the board. At that point I didn't want to significantly change the layout so I went with stabilisation values that worked solidly.
The circuit is however a simple voltage follower at DC, with the output at half-rail with fine trim to cancel the opamp input offset voltages. The 1M resistor also matches the one on the inverting input of the 'measuring' opamp which hopefully also minimises bias current offsets (this does seem to work as output offset at the uV level is maintained pretty well over normal ambient variations).
P.S. @bdivi, I hadn't noticed your post with the implementation of the "Gyro picometer" (undeserved fame at last!
). It looks a neat implementation and it's good to see that it performs so well.
The LMC662 was 'twitchy' in driving the ground (low frequency instability). I suspect that it is because 'ground' in my case was the copper groundplane of the board.
I'd guess that 'ground' was one plate of a 10-20 pF capacitor depending on the size of your board.
I think Gyro's mistake was missing that 'ground' was just another wire/net/node from the electrons' point of view. I find it helpful in schematics for this kind of setup to use a symbol* for an equipotential point that isn't the classic earth or chassis symbol and separately show a direct connection to ground from that symbol to reinforce the point that there is nothing magical about the 'ground' connection, that it's just another node.
* Can't be bothered to generate and upload a picture. I usually use the inverted hollow triangle with a single letter designator (A and D being the most common).
I think Gyro's mistake was missing that 'ground' was just another wire/net/node from the electrons' point of view.
Hmm, maybe, maybe not. In this case that node is 'ground', It is connected to the board copper plane, the BNC connector outer and the metal case itself. In this case the ground symbol is rather appropriate. It's the battery terminals that are being 'positioned' by the opamp to span each side 'ground'. There shouldn't be any assumption that either of the battery terminals is a ground reference.
EDIT: On reflection a 'chassis' symbol would be more appropriate but I don't think that would be any less confusing than the ground symbol. It is still the node that will be grounded externally (via the ground terminal) during use.
Edit 2: Just looking back at the schematic, I did actually use the chassis symbol.
1G @ 10hz = 13fA rms resistor noise, approximately? That would be not including current and 1/f noise, correct?
Thanks
If you sample at 10Hz the bandwidth should be about 5Hz and RMS thermal noise current of the resistor would be about 9fA at room temperature. To reduce it you need to use a larger resistance. A 100G resistor would give you ~ 0.9fA RMS noise. Amplifier noise components are added to that figure, though good electrometer-grade opamps add very little.
Cheers
Alex
The noise bandwidth will be determined by the anti-alias filter; without one it will be the bandwidth of the amplifier and the analogue bandwidth of the ADC - which for a 1MSPS convertor (not uncommon on a micro-controller) could be 10, 20 or even 30MHz+. (All the noise above the Nyquist rate will fold down into the passband). By sampling much faster than 10Hz, a simpler anti-alias filter can be used with a digital low pass filter removing noise above 5Hz.
Can someone explain for a newb why series connection of lower value resistors are nogo?? I Something related to noise?? Dielectric absorption is too high because of the "end cap" of leads and there is a series of voltage sources because of DA??
I happen to have a pair of OPA111AMs from B&B (salvaged board). Are they any good for this kind of job? Electronics aren't my area of specialty so I stuggle with the datasheets, but to me it seems that the low as possible input bias current is what you are after and what matters here and since that current is for OPA111AM a pA range (sub) it will be nogo, since it should be on femtoAmp range? Or is it just matter of complexity of balancing the input that goes beyond DIY.
The OPA111 is a great choice down into the picoamp range but its performance will be limited by its change in bias current with temperature which doubles roughly every 10C.
Can someone explain for a newb why series connection of lower value resistors are nogo?? I Something related to noise?? Dielectric absorption is too high because of the "end cap" of leads and there is a series of voltage sources because of DA??
I can't immediately see where the series connection being 'nogo' reference comes from. Series connection is quite often used in high voltage dividers, it spreads a voltage drop across multiple packages, increasing breakdown voltage and it can actually reduce shunt capacitance caused by the packages (both effects really coming from you effectively building a physically longer resistor). Of course the devil is in the detail - how you construct the chain, stray leakages if you construct on a PCB etc. I don't think dielectric absorbtion comes into the equation though.
Can someone explain for a newb why series connection of lower value resistors are nogo?? I Something related to noise?? Dielectric absorption is too high because of the "end cap" of leads and there is a series of voltage sources because of DA??
I can't immediately see where the series connection being 'nogo' reference comes from. Series connection is quite often used in high voltage dividers, it spreads a voltage drop across multiple packages, increasing breakdown voltage and it can actually reduce shunt capacitance caused by the packages (both effects really coming from you effectively building a physically longer resistor). Of course the devil is in the detail - how you construct the chain, stray leakages if you construct on a PCB etc. I don't think dielectric absorbtion comes into the equation though.
I think I misunderstood somepoint of this thread, now I rereaded it through I don't know where I got such idea, but the subject goes over my (currently)head. Maybe I mimick this design and try to salvage those ($50 how on earth they are so expensive) OPA111s from that board and test if I can get to under nA range (without anykind of true accuracy).
I made a quick calculations and it seems that Gohm range of resistor the price is pretty constant 35 to 55 euros per 10GOhm so the series idea wouldn't be a such economically practical solution anyway even with 1000 SMDs
For mechanical construction firts thing that came to my mind were PTFE poles and zigzag air network of resistors between them.
The OPA111 is a great choice down into the picoamp range but its performance will be limited by its change in bias current with temperature which doubles roughly every 10C.
That's going to be an issue with anything with a JFET input. The only way to avoid that is to go with MOSFETs and down in the pA ranges the lower noise of a JFET is going to be a winner every time compared to MOSFETs, plus there's a tendency for MOSFET inputs to have a higher 1/f noise corner.
For instance, OPA111 (JFET, 800 fA typical bias) 8 nV/sqrt(Hz) @ 1 kHz, LMC662 (MOSFET, 2fA typical bias!) 22 nv/sqrt(Hz) @ 1 kHz. The LMC662 data sheet doesn't tabulate any noise figures other than @ 1kHz and the graphs show a very poorly defined 1/f corner such that noise is still falling at 10 kHz, the OPA111 has a well defined, if high, noise corner at around 1kHz (The more modern OPA627 has a much more respectable 100 Hz 1/f noise corner).
I've been toying with the idea of building this circuit for years-- even have the components stored somewhere. One improvement I was thinking about was changing the mongo resistor for a tee network. Would it be advisable in this case?
The OPA111 is a great choice down into the picoamp range but its performance will be limited by its change in bias current with temperature which doubles roughly every 10C.
That's going to be an issue with anything with a JFET input. The only way to avoid that is to go with MOSFETs and down in the pA ranges the lower noise of a JFET is going to be a winner every time compared to MOSFETs, plus there's a tendency for MOSFET inputs to have a higher 1/f noise corner.
For instance, OPA111 (JFET, 800 fA typical bias) 8 nV/sqrt(Hz) @ 1 kHz, LMC662 (MOSFET, 2fA typical bias!) 22 nv/sqrt(Hz) @ 1 kHz. The LMC662 data sheet doesn't tabulate any noise figures other than @ 1kHz and the graphs show a very poorly defined 1/f corner such that noise is still falling at 10 kHz, the OPA111 has a well defined, if high, noise corner at around 1kHz (The more modern OPA627 has a much more respectable 100 Hz 1/f noise corner).
That is exactly why I have used graded LMC6081s (practically the same as the LMC662 but with better precision) in the past when I needed lower input bias current than a JFET input although there *are* some JFET input operational amplifiers like the AD549 with a bias current significantly below that of the OPA111.
At high temperatures, bipolar input operational amplifiers become competitive.
So as newb with a straight thinking, without much considerations of realities.
A deep frozen MOSFET input is the best? (in reality moisture and temp control would be a nightmare)
Is the time driven through those classic discrete multistage darlington and FET-BJT hybrids etc. amplifier designs, when measuring nothingness.
For my calculation of the cost of gigaohm resistors I used the easiest source for me (as non-professional, non institutional, non commercial, non-corporate buyer) it is farnel.uk (since my local electron shed no more takes custom orders to other as used to), so it is obviously not universally applicable. I'm going through my buckets of old stuff I hoarded as a kid, but I think I'm lucky if I find anything over 1GOhm and the condition of such parts is random.
