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Hi, I'm using a H11F1 photo fet optocoupler for isolated resistance control (the H11F1 does 100R to 300meg)
and i'm driving the opto LED through a filter from a 5V MCU PWM based DAC (30kHz).
The filter is two RC filters in a row. Both using 100R and 100uF. (The 100R also function as the LED current limit).
If i go any lower than 100uF for the capacitors i can start to detect some of the PWM DAC noise getting through the opto and changing its output resistance.
The resistors are somewhat fixed due to needing the current to run the opto LED.
The product works fine as is, but I would really like to use less than 100uF capacitors, it seems rather excessive.
Does anyone have any better ideas to filter this sort of thing. Maybe a few LC filters or a RC then a LC would work better? what about LC etc..
I'm just not very knowledgeable about the type of passive filters and what works better in what situation.
I could do an active filter but i would prefer not to add any more ICs to the BOM. (Unless you tell me it will work MUCH better in this situation)
Any ideas?
Thanks.
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The first problem here is C16. At powerup the capacitor is quickly charged by the poor LED, with no current limitation.
I would drive the LED using a bipolar transistor, linearizing the current with a resistor on the emitter. I would then drive the transistor base via an RC filter with way smaller C and way higher R values. Full flexibility regarding NPN/PNP choice, inverted logic, etc. We can easily switch to an LC if we want a steeper filter response.
Could this be viable in your project?
If you give more details regarding the current range through the LED, we can define some values.
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The first problem here is C16. At powerup the capacitor is quickly charged by the poor LED, with no current limitation.
That is a very good point. I will fix that in next version. Thanks
I would drive the LED using a bipolar transistor, linearizing the current with a resistor on the emitter. I would then drive the transistor base via an RC filter with way smaller C and way higher R values. Full flexibility regarding NPN/PNP choice, inverted logic, etc. We can easily switch to an LC if we want a steeper filter response.
Could this be viable in your project?
If you give more details regarding the current range through the LED, we can define some values.
Anything is viable for the project, but simple and low part count is better.
Currently the IR LEDs are being driving at 18mA. but 15-20mA should be fine.
Ideally it would be nice if i could drive the LED in way that was more linear.
Currently the response between PWM and resistance is very non-linear which mean i don't have very fine control at one end. But that is a separate issue. -
Is there a strong need to filter a diode input current? Isn't it easier to filter optocoupler output?
There is an another way to filter - use active opamp integrator instead of passive RCRC.
If optocoupler non-linearity makes inconvenience there is a way to negotiate it with two identical optocouplers and opamps. I can't find right example now. If you go this way then you can make an active filter (active integrator) simultaneously.
Oh, here it is. Use a photoelectric-FET optocoupler as a linear voltage-controlled potentiometer. -
The attached schematic offers a good view of the components needed ando also a simulation of the waveforms.
In red the diode current, with no appreciable ripple.
You can adjust the values to match the desired dynamic.
There is no turn-on current rush either. The response is also nicely linear thanks to the emitter resistor.
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A much better functionality can be achieved by keeping the BC547 and removing R4 and C2. Instead add a series RC link from collector to base. Also a collector resistor.
This will give a true 2nd order response and integration.
You'll need to do a bit of maths here, including playing with R1 and R2
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Is there a strong need to filter a diode input current? Isn't it easier to filter optocoupler output?
There is an another way to filter - use active opamp integrator instead of passive RCRC.
If optocoupler non-linearity makes inconvenience there is a way to negotiate it with two identical optocouplers and opamps. I can't find right example now. If you go this way then you can make an active filter (active integrator) simultaneously.
Oh, here it is. Use a photoelectric-FET optocoupler as a linear voltage-controlled potentiometer.
Thanks for that! -
The attached schematic offers a good view of the components needed ando also a simulation of the waveforms.
(Attachment Link)
(Attachment Link)
In red the diode current, with no appreciable ripple.
You can adjust the values to match the desired dynamic.
There is no turn-on current rush either. The response is also nicely linear thanks to the emitter resistor.
I will build this up and give it a try.
Thanks. -
1. As others have said, C16 is in parallel with the LED's dynamic impedance, which is quite low, so at the very least it has to be many times larger than shown to get the expected pole. The resistors also act in parallel with C14, raising its pole -- when you have a chain of RC filters, you cannot get repeated poles (let alone complex poles), instead the poles push each other away and you end up with a very, very soft response.
The best you can do is a tapered impedance, say R6 is 10 ohms, R8 is 220 and say we add another resistor between C16 and the LED of 220 ohms as well. This gives C16 about 100 ohms around it, and C14 around 10 ohms; if we set C14 = 1mF and C16 = 100uF, we get similar poles, and about as sharp a cutoff as we can expect.
2. You're also coupling maximum supply ripple into the poor LED. This was hinted at above as inrush, but perhaps not made fully explicit. Every minute variation between DGND and VCC5 is coupled directly into the LED.
Trivial fix: wire the LED to DGND, or ground the caps to VCC5. Keep ground references consistent!
3. As has also been mentioned, active filters don't need to be difficult. A single transistor will do, though it's not like it's a pain to put in a SOT-23-5 opamp, say a TLV2371. Most of the layout space, and assembly cost, will go to passives regardless.
Just to show it works in any technology, here's one I designed for audio frequencies using a vacuum FET:
Note the response shelves at HF (to a ~ -48dB asymptote), which is due to the finite gain of the active device (about 5mS). You could use, essentially the same circuit with a BJT, though you'll want the resistor values smaller to adjust for a 5V supply, as I'm guessing you won't have 100V available.
Also easy to put into a simulator, so you can tweak the values for desired response and availability. (Seems I happened to get best results with the 6.8n caps I had on hand, among other things, so that worked out nicely there.) The R and C time constants are a bit peculiar, due to the gain affecting them -- indeed, it is possible to make an oscillator here (though that requires one or more additional RC stages to the grid/base), so just start by scaling everything linearly, then tweak parts by say 20% at a time until it's where you want it.
You want lowpass, so obviously C1, C2, C8, R1 and R3 go away, and probably R6 and R7 as well.
Note that the result is almost identical to brabus's circuit, and this is no accident. The two lessons we can apply are these: a. the R and C values are equal, so the cutoff will be very soft, much softer than needed, which means attenuation or response time is left on the table; b. by moving C2's ground reference to the emitter instead, we get the Sallen-Key topology where we can peak response for best risetime or sharpest attenuation. We also know we can get a 3rd pole for free, by putting an RC in front of the filter (of course again changing R and C values to suit).
Noteworthy, the open-collector output has a constant-current characteristic, so any VCC5 supply ripple is dropped primarily across the transistor, with very little effect on LED current. This serves the same function as changing LED or cap ground reference above.
The LED can also be in the emitter circuit (in which case the collector goes straight to VCC5), the downside is a higher offset voltage (Vf + Vbe) which wastes more PWM% range. In this configuration you are still limited to some maximum PWM less than 100%, for a similar reason (actually Vf + Vce(sat) - Vbe, more or less), but it is a wider range, so it's a good choice.
4. Final note -- whatever the PWM is coming from, will influence the output just the same. That is, if it's a microcontroller running from VCC5, then any variation in VCC5 will again act, proportionally, on the output. A PWMDAC is a multiplying DAC, and the output equals PWM% times VCC. The PWM is well filtered, so you're safe from high frequency variation, but low frequency variation, drift, whatever, is straight through. If greater stability is desired, buffer the PWM signal with a logic gate (preferably an SPDT analog switch) supplied from a stable VREF.
Note that the transistor does introduce a source of drift, as Vbe depends on temperature. This can be addressed with a second transistor in a zero-offset follower (it's not actually zero, but the Vbe's and tempcos cancel out a heck of a lot better than no cancellation whatsoever):
At this point you're easily spending as much effort as plunking down an op-amp, so weigh your options accordingly.
Ed: oh yeah, I do have an even more directly relevant example--
https://www.seventransistorlabs.com/Images/EE409_Schem_r_4.png
note the two PWM channels, buffered by gates to VREF, and filtered. This gave a precision output for a digitally controlled bench SMPS.
Tim -
<Lots of good info>
Tim
Wow. Thanks for that. Lots of useful info there for me to digest.